Spectrally Efficient Long-Haul Optical Networking Using 112-Gb/s Polarization-Multiplexed 16-QAM

10 Pages·1.668 MB·English

by  Winzer P.J., Gnauck A.H. et al.

JOURNALOFLIGHTWAVETECHNOLOGY,VOL.28,NO.4,FEBRUARY15,2010 547
Spectrally Efficient Long-Haul Optical Networking
Using 112-Gb/s Polarization-Multiplexed 16-QAM
P.J. Winzer, A. H.Gnauck, C.R. Doerr, M.Magarini,and L.L.Buhl
Abstract—Wediscussthegeneration,wavelength-division-mul- at fixed signal bandwidth. This process started with dif-
tiplexed (WDM) long-haul transmission, and coherent detection ferential quadrature phase shift keying (DQPSK) [3], which
of 112-Gb/s polarization-division-multiplexed (PDM) 16-ary
allowed point-to-point SEs of up to 1.6 b/s/Hz [4], and com-
quadrature amplitude modulation (16-QAM) at a linerate of14
fortably enabled optically-routed networking at 40 Gb/s on
Gbaud and spectral efficiencies beyond 4 b/s/Hz. We describe
the (off-line) digital signal processing and blind filter adaptation a 50-GHz WDM grid (SE of 0.8 b/s/Hz), including multiple
algorithms used in our intradyne receiver and characterize its passages though reconfigurable optical add/drop multiplexers
performanceusingbothsimulatedandmeasured16-QAMwave- (ROADMs) [5]. In a next step, polarization division multi-
forms.Wemeasurearequiredopticalsignal-to-noiseratioof20.2
plexing (PDM) was added to boost the SE to 3.2 b/s/Hz in
dB(0.1-nmreferencebandwidth;(cid:0)(cid:2) (cid:0)bit-errorratio),3.2-dBoff
point-to-point applications using direct detection [4], and to
thetheoreticallimit.Westudytheeffectsoffiniteanalog-to-digital
converter resolution, laser frequency offset, laser phase noise, 2 b/s/Hz in an optically-routed environment using coherent
and narrowbandopticalfiltering. Ourexperimentson a 25-GHz detection [6]. Today, PDM-QPSK is considered an attractive
WDM grid (4.1-b/s/Hz spectral efficiency) reveal a 1-dB penalty optionfor100-Gb/sopticallyroutedtransportsystems.
after 7 passes though reconfigurable optical add/drop multi-
Furthercompressionofthesignalspectrumrequireshigher-
plexers(ROADMs)andanachievabletransmissionreachof1022
level( -ary)modulationformats,beneficiallyincombination
km of uncompensated standard single-mode fiber. At a spectral
efficiency of 6.2 b/s/Hz (16.67-GHz WDM channel spacing) a withcoherentdetection,withanSElimitedto2 using
transmissionreachof630kmisattained. PDM. In practice, and assuming optically-routed networking
Index Terms—100G Ethernet, coherent detection, optical net- with multiple state-of-the-art ROADMs, 50% of this value,
working,opticaltransmission,quadratureamplitudemodulation i.e., maybeobtained.Furthermore,higherSEscomeat
(QAM),wavelength-divisionmultiplexing(WDM). theexpenseofincreasinglymorestringentsignal-to-noise-ratio
(SNR)requirements.ThisisvisualizedinFig.1[7],wherere-
centexperimentalresultsforquadratureamplitudemodulation
I. INTRODUCTION
(QAM)arecontrastedwiththeoreticallimitsforthelinearcom-
E NABLEDbywavelength-divisionmultiplexing(WDM),
munication channel. The net impact of higher SNR require-
fiber-optic transport capacities havebeen growing expo-
ments on the achievable transmission reach depends on fiber
nentiallyoverthelast20years,justenoughtomeettheannual
type, amplification scheme, and impact of fiber nonlinearities
growth in data traffic of several tens of percent per year gen-
[8], [9]. At 1 Gbaud, constellations as high as 128-QAM with
eratedbyourmulti-mediacommunicationandinformationso-
aspectralefficiencyof9.3b/s/Hz(includingtherequiredover-
ciety[1].Atafixedopticalamplificationbandwidth,increasing
headforforwarderrorcorrection,FEC)havebeendemonstrated
the transmission capacity requires increasing the spectral effi-
[10].At56Gb/s,aspectralefficiencyof7.0b/s/Hzwasobtained
ciency(SE),i.e.,thenetper-channelbitrate dividedbythe
usingPDM32-QAMandorthogonalfrequency-divisionmulti-
WDMchannelspacing
plexing(OFDM)[11].Noteinthiscontextthatmulti-tonemod-
ulation(OFDMorcoherentWDM)hasthesameSEandSNR
(1)
limitsassingle-carriermodulation.At114Gb/s,8-PSK[12]and
8-QAM[13]haveachievedSEsof4.2b/s/Hzforpoint-to-point
Up until a few years ago, virtually all optical transmission
applications(noROADMs).
systems employed binary intensity or phase modulation [2],
Inthispaper,wereportonaseriesofexperimentsconducted
and the required increase in SE was satisfied by advances in
at 112 Gb/s using PDM 16-QAM at 14 Gbaud, expanding
high-speed electronics (higher ) as well as in laser- and
uponourinitialreportsin[14]–[16].Wedemonstratelong-haul
optical-filter frequency stability (reduced ). In the early
optically-routed networking on a 25-GHz WDM grid, and
2000s, the speed of opto-electronic modulation and detec-
point-to-point transmissionon a 16.6-GHzgrid, yieldinga SE
tion approached the bandwidths of stable optical filters, and
of 6.2 b/s/Hz, the highest reported for single-carrier 112-Gb/s
higher-order modulation formats had to be used to increase
signals.
ManuscriptreceivedJune26,2009.FirstpublishedSeptember09,2009;cur- II. 16-QAMTRANSMITTER
rentversionpublishedFebruary01,2010.
P.J.Winzer,A.H.Gnauck,C.R.Doerr,andL.L.BuhlarewithBellLabo- The setup of our 16-QAM transmitter is shown in Fig. 2.
ratories,Alcatel-Lucent,Holmdel,NJ07733USA(e-mail:peter.winzer@ieee. First, four copies of a true 14-Gb/s binary pseudo-random bit
org).
sequence(PRBS)oflength aregeneratedandamplified
M.MagariniiswiththePolitecnicodiMilano,20133Milan,Italy.
DigitalObjectIdentifier10.1109/JLT.2009.2031922 using high-speed electrical driver amplifiers. Although these
0733-8724/$26.00©2010IEEE
548 JOURNALOFLIGHTWAVETECHNOLOGY,VOL.28,NO.4,FEBRUARY15,2010
Fig.1. SpectralefficienciesachievableasafunctionofrequiredSNRperbit,
referencedtoasinglepolarization.TheShannonlimitappliestoidealcoding
andmodulation.ThesymbolsassumeQAMandenhancedFECwith7%over-
headanda(cid:0)(cid:0)(cid:2)(cid:3) correctionthreshold.Circlesaretheoreticallimits,squares Fig.3. Setupofthepolarization-diversitycoherentreceiver.
representmeasuredpointsatmulti-Gb/slinerates.
III. COHERENTRECEIVERHARDWAREANDALGORITHMS
A. Opto-ElectronicReceiverFront-End
The opto-electronic front-end of our intradyne receiver is
showninFig.3.Thesignaliscombinedwithalocaloscillator
(LO) laser, typically the same type of ECL as used for the
signal under test, in a polarization-diversity 90-degree optical
hybrid.TheLOistunedtowithinapproximately MHzof
the received signal’s center frequency. The four pairs of bal-
ancedoutputsfromthehybridaresampledandasynchronously
digitized at 50 GSamples/s using a commercial 4-channel
real-time oscilloscope, with a nominal 8-bit resolution of the
analog-to-digital converters (ADCs) and a frequency-depen-
dent effective number of bits (ENoB) between 4 and 5. All
Fig.2. Setupofthe16-QAMtransmitter. results shown here are based on offline processing of at least
onemillionsignalsamplesoneachofthefoursamplestreams,
corresponding to more than 250 received symbols ( 2
signals are still binary, signal degradations such as a reduced millionbits)forbit-errorcountingaftertheconvergenceofall
risetimesorovershootshavetobeavoidedinordertokeepim- receiverloops.
pairmentsofthegeneratedfour-levelsignaltoaminimum.For
B. ReceiverDigitalSignalProcessing
each quadrature, one binary drive bit stream is interpreted as
the most significant bit (MSB), and one is attenuated by 6 dB AsshowninFig.4,thefoursamplestreamsfromtheADCs
andinterpretedastheleastsignificantbit(LSB)ofa2-bitdig- areinterpretedasrealandimaginaryparts(IandQ)oftwocom-
ital-to-analogconverter(DAC).MSBandLSBstreamsarethen plexsamplestreams,oneforeachpolarization.Inafirststep,the
resistivelycombinedwithamultiple-bitdecorrelationdelay(12 algorithmperformsaccouplingandcorrectsforspecific(static)
and 18 bits, respectively) as shown in Fig. 2. An eye diagram opticalfront-enderrors,suchasasamplingskewbetweenIand
of the four-level signals making up in-phase (I) and quadra- Q signal components orphase errorswithin the 90-degree hy-
ture (Q) components of the 16-QAM signal is also shown in brid.Then,digitalantialiasingfilteringisperformed.Theexact
the figure. The peak-to-peak voltage of the four-level signals antialiasingfiltershapehaslittleimpactontheperformanceof
is approximately 3.5 V. In some of our experiments,the drive thereceiver;rectangularandGaussianfiltershapeswereused.
signalsarelow-passfiltered(LPF)toelectricallysuppressmod- Then,abulkchromaticdispersion(CD)filtercompensatesfor
ulationsidelobesbeforebeingappliedtoanintegratedLiNbO dispersion of the transmission line. This linear filter is imple-
double-nested Mach–Zehnder (I/Q) modulator. The 16-QAM mented in the frequency domain using fast Fourier transforms
opticalintensityeyediagramwithitsthreeintensitylevelscor- (FFT)andmultiplicationwiththequadraticspectralphasechar-
responding to the three rings that make up a square 16-QAM acteristicofCD.Subsequently,theclockfrequencyisrecovered
constellationisalsoshowninFig.2. bytakingtheFFTofthesignalpowerwaveformanddetecting
Asalightsource,wetypicallyuseaC-bandtunableexternal the pronounced 14-GHz tone. Using the recovered clock fre-
cavitylaser(ECL)withameasuredlinewidthof 100kHzfor quency,thesignalisdown-sampledtoasynchronous28GSam-
thechannelundertest.Interferingchannelsusedistributedfeed- ples/s(2xoversamplingat Gbaud),albeitwithayet
back(DFB)laserswithlinewidthsof 2MHz.Weemploytwo unknownclock phase. Then, the detection algorithm performs
different16-QAMtransmittersofthesamestructure,oneforthe thefollowingtasks:
setof‘even’andoneforthesetof‘odd’WDMchannels.Both • Sourceseparationtoadaptivelyrestoretheoriginal and
setsareindividuallymodulatedbeforebeingcombinedandpo- polarizations of the transmit signal from the randomly
larization-multiplexed,asdetailedinSectionV. rotatedreceivesignalpolarizations.
WINZERetal.:SPECTRALLYEFFICIENTLONG-HAULOPTICALNETWORKING 549
fromasinglecircleofradius inthecomplexplane(‘constant
modulus’).Thefiltercoefficientsareadaptedaccordingto
(3)
which minimizes . Here, stands for either
or and denotes the th filter tap of any one
of the four FIR filters; denotes the (complex conjugate)
signal at the equalizer input at time step – , and is a
convergence parameter, typically chosen on the order of
for our receiver. CMA adaptation works particularly well for
Fig.4. Blockdiagramofthedigitalsignalprocessingusedwithinthecoherent (single-ring) PSK constellations, where it is often the only
receiver.
adaptation algorithm used [20]. For QAM constellations,
which are generally composed of multiple rings (cf. inset to
Fig. 2), the error in Eq. (2) will not be reduced to zero by the
• Adaptive equalization to counter randomly varying
CMA; nevertheless, minimizing the error adjusts the filter
channelimpairments.
taps such that the equalized constellation becomes reasonably
• Recoveryofthecorrectsamplingphase.
compact, which yields sufficient pre-convergence based
• Frequencyrecoverytocompensatefortheresidualrotation
on which one can safely switch to decision-directed (DD)
oftheequalizedconstellationdiagramduetothenon-zero
equalization (see [17]–[19] for a discussion in single-source
intermediate frequency (IF), i.e., the beat frequency be-
(non-PDM) situations). Several modifications to the CMA
tweensignalandLOlaser.
have also been proposed [21], [22], including radius-directed
• Phase recovery to keep the square QAM constellation
(‘multi-modulus’,MMA)decision-aidedalgorithms[23]–[26],
aligned with the fixed rectangular decision boundaries
whereonefirstmakesadecisionontheringareceivedsymbol
(rectilineargridintheinsettoFig.2),whichareoptimum
most likely belongs to and then adapts the equalizer using a
for QAM detection in the presence of circularly sym-
CMA based on knowledge of the correct ring radius , i.e.,
metric noise. Note that in the context of synchronization
one uses the error . One drawback of
techniques for digital receivers, frequency and phase
this approach is the fact that it relies on the correct decisions
estimation are often treated together under the name of
regardingthetransmittedringradius;sincetheringspacingin
carrierphaserecovery.
QAM is generally smaller than the minimum symbol spacing
The following subsections describe these functional blocks in
(0.87 and 0.54 for square 16-QAM), these decisions
detail.
show a significant number of errors for heavy noise loading
and/or for severe signal distortions; in agreement with [27],
we observed a reduced robustness in filter pre-convergence
C. AdaptiveSourceSeparationandEqualization
using the multi-ring CMA and therefore opted for the classic
Adaptive source separation, equalization, and sampling CMA. As a default starting condition for our regular CMA,
phase recovery are simultaneously performed by the lattice we use unit impulses for and (
filter with transfer functions , and for and zero otherwise) and all zeros
shown in Fig. 4, which represents the frequency-de- for and , and only change this starting condition if
pendent inverse Jones matrix of the transmission channel. pre-convergenceisnotobtained.
Eachfilterblockisimplementedinthetimedomainasafinite After CMA pre-convergence, an initial estimate of carrier
impulseresponse(FIR)filterwith fractionally-spaced frequency offset and symbol phase was obtained as detailed
(T/2) taps, sufficient to allow for reliable operation under a in Section III.D, and filter adaptation was switched to deci-
pulsebroadeningofupto 3symbols[17]. sion-directed (DD) operation. However, instead of using stan-
In the context of this paper, we are interested in blind filter dardleast-meansquares(LMS)adaptationwithanerrorsignal
adaptation, i.e., adaptation without the use of a training se- , where is the correct angular
quence.Inafirststep,weusethewell-provenconstantmodulus back-rotationfollowingfrequencyandphasetracking,weused
algorithm (CMA), a blind filter adaptation algorithm that is thephase-independenterrorsignal[28]
simple,robust,andworksindependentofcarrierfrequencyand
(5)
phase, both of which are initially not available to the receiver
[17]–[19]. In essence, the CMA minimizes the time-averaged
togetherwiththecorrespondingfilterupdatealgorithm
error
(6)
(2)
where is typically chosen on the order of . This filter
reflecting the average distance of the equalized received sym- adaptationisinterleavedwithadecision-directedphase-locked
bols ( beingthesymboltimeindex)attheoutputofthefilter loop (PLL), as shown in Fig. 4. Note that the decisions are
550 JOURNALOFLIGHTWAVETECHNOLOGY,VOL.28,NO.4,FEBRUARY15,2010
basedonthe(optimum)standardrectilineargridofQAMdeci-
sionboundariesasopposedtoradialdecisionboundaries[23],
[24]; only the error signal is exclusively based on radial in-
formation,whichdecouplestheequalizerupdatefromresidual
phasetrackingerrorswithinthePLL.
Throughout our experiments, we employed Gray-coded
16-QAMwithoutdifferentialpre-coding.Withintheparameter
spacestudiedinthispaper,wedidnotencounteranycycleslips Fig. 5. (a) Decision-directed PLL; (b) In acquisition mode, a max-
thatcouldleadtolongerrorburstswhendifferentialpre-coding imum-likelihood estimate is made with respect to the unknown rotation
ofthereceivedconstellation.
isnotused[29],[30].Beforeoutputtingvaliddata,thereceiver
hastotestagainstashortknownreferencepatterntodetermine
thecorrectrotationoftheQAMconstellation.Thisreferencing
processatthereceiverisregularlydoneinsystemsemploying
advancedmodulationformats,evenforsimpledirect-detection
systems using differential phase shift keying (DPSK), where
thereceiverhastocheckforapossiblepatterninversion.
D. FrequencyandPhaseEstimation
AfterCMApre-convergencethereceivedsignalisreasonably
welldecomposedintoitstwopolarizationsandisroughlyequal-
izedwithineachpolarization,buttheconstellationisstillspin-
ningattheIF,iswigglingduetolaserphasenoise,andisrotated
toarandomphaseanglecomparedtotherectilineardesiredde-
cision boundaries that are aligned with the real and imaginary
Fig.6. Back-to-backBERmeasurementsofourinitialsetup.
axes.Inordertoachievefrequencylockandphasealignment,a
standarddecision-directedPLL[19]isimplementedonablock
of 1000 symbols, as shown in Fig. 5(a). In contrast to com-
signal-to-noiseratio(OSNR,noisereferencedtoa0.1-nmreso-
monlyemployed loopfilters, ourloopfilteris implementedas
lutionbandwidthandbothpolarizations)of18dBtoachievea
arunningaverageover,respectively, and samplesofthe
BERof ,whichcorrespondstoanimplementationpenalty
phase difference between the pre-decision and post-deci-
ofabout4dBcomparedtothetheoreticallimit[31].
sionsignalsamples,andtheweightedoutput(weightingfactor
PolarizationmultiplexingrequiresanOSNRof22dBforthe
) of the two integrators is used as the input to the
measuredQAMwaveforms,revealinganexcesspenaltyof1dB
digitalvoltage-controlledoscillator(VCO).Theparameter is
forPDM.Therecovered16-QAMsignalconstellationsforboth
typically chosen around 0.95. The integration time constants
polarizations at 23-dB OSNR are shown as an inset to Fig. 6.
are chosen on the order of 10 for phase estimation and
Subsequentimprovementstobothtransmitterandreceiverwere
1000forfrequencyestimation.Inordertofindthecorrect
abletoreducethePDMexcesspenaltytolessthan0.2dB.
phaseangle,westepthrough10referenceconstellations,rotated
inincrementsof ,andpicktherotationangleoftherefer-
B. ParameterSensitivities
enceconstellationthatshowsthesmallestmean-squareerror(cf.
Fig. 5(b)). Thisprocess corresponds toa maximum likelihood In order to understand the sensitivity of our algorithms to
(ML) search for the correct phase angle while simultaneously various system parameters, we performed extensive Monte
estimatingtheIF.Wefoundthisdecision-directedapproachto Carlo simulations of single-polarization coherent receiverper-
workmuchmorereliablythanmulti-modulusvariationsofthe formance. The 16-QAM waveform used for these simulations
Viterbi-Viterbialgorithm[19],[26],forthesamereasonsasdis- is based on 64 repetitions of a random 16-QAM sequence
cussedinSectionIII.CalongwiththeCMAandDDfilteradap- of length 2048, oversampled by a factor of 8. The employed
tation algorithms. After initial frequency and phase lock, the modulator structure, including band-limited drive waveforms,
decision-directedPLLcontinuesitsoperation,interleavedwith is replicated by the simulations. The intensity eye diagram of
DDfilteradaptation. the simulated waveform as well as the symbol constellation
(including the symbol transition paths in the complex optical
fieldplane)isshownintheinsettoFig.7(a).Weusethesame
IV. BACK-TO-BACKCHARACTERIZATION
algorithmsforsimulationasusedinourexperiments,andarrive
at a simulated required OSNR of 14 dB at ,
A. RequiredOpticalSignal-to-NoiseRatio(OSNR)
which corresponds to the theoretical limit and hence proves
Fig. 6 shows back-to-back single-channel bit-error-ratio the absence of purely algorithm-induced sensitivity penalties
(BER) measurements of the 16-QAM signal of our initial inoursetup.
setup [14], without electrical pre-filtering of the modulator ReceiverperformanceasafunctionoftheADCresolutionis
drivesignals.Thesingle-polarizationsignalrequiresanoptical shown in Fig. 7(a). Between 4 and 5 bits are needed to avoid
WINZERetal.:SPECTRALLYEFFICIENTLONG-HAULOPTICALNETWORKING 551
Fig.8. Toleranceofthesystemtonarrowbandopticalfiltering.Symbolsare
experimentalmeasurements.Shadedareasaresimulationsfor1st,2nd,and4th
ordersuper-Gaussianfilters,boundedbythetwoextremecaseswherefiltering
takesplacebeforenoiseloadingandafternoiseloading,respectively.
of1st,2nd,and4thorder,withthenormalized3-dBfilterband-
width shown on the x-axis. Depending on whether the OSNR
degradation takes place before or after the narrowband filter,
Fig.7. (a)SimulatedOSNRpenaltyasafunctionoftheADCresolution;inset:
differentresultsareobtained.(Inpractice,noisewillbeadded
intensityeyediagramandsymbolconstellationwithsymboltransitionsofsim-
ulated16-QAMwaveform;(b)Curve:SimulatedOSNRpenaltyasafunction bothbeforeandaftertheseveralfilterspresentinasystem, so
oftheaggregatelaserlinewidth(signalplusLO);symbols:measuredOSNR thattheactualperformanceofanoptically-routedsystemisex-
penaltyusinglaserswithdifferentlevelsofphasenoise.
pectedtofallinbetweenthesetwoextremes).Ingeneral,worse
results are obtained when introducing noise after narrowband
filtering,sinceinthiscasethestrongestnoise-enhancementef-
penaltiesinexcessof1dB.Thesevaluesareslightlymorefa-
fectduetotheadaptiveequalizerwithinthecoherentreceiveris
vorablethanthosereportedin[29].Recallthattheeffectiveres-
observed.NoiseenhancementisnotobservedwhentheOSNR
olutionprovidedbythereal-timeoscilloscopeisbetween4and
is degraded before narrowband filtering, since in this case the
5 bits.
filter acts on both signal and noise and the equalizer re-estab-
Regardingthelock-inrangeofourfrequencyrecoveryloop,
lishes white noise conditions while simultaneously equalizing
oursimulationsshownegligiblereceiverperformancedegrada-
the signal. The symbols are experimental measurements using
tionsforvaluesoftheIFupto 100MHz,alsoforsubstantial
two50GHz/100GHzinterleaversthatweredetunedrelativeto
opticalnoiseloadingandtypicallaserlinewidthsof 100kHz.
eachother,therebyemulatingnarrowopticalfiltersof8,11,18,
Inourexperiments,wekepttheIFbelow 20MHz,whichal-
and26GHzbandwidth,albeitwithvaryingfiltershapes.Noise
waysyieldedreliablefrequencylocking,eveninthepresenceof
loadingwasperformedafterfilteringintheseexperiments,and
severenoiseloadingandsignaldistortions.
a reasonable agreement between measurement and simulation
System performance as a function of the overall laser
isobserved.
linewidth(signalplusLO)isquantifiedinFig.7(b).Thecurve
showssomestatisticaluncertaintyduetotheunderlyingMonte
V. WDMEXPERIMENTS
Carlo simulations. The sensitivity to laser phase noise closely
matches the results of [30] but shows better performance than ForallourWDMexperimentsweusedtheexperimentalsetup
reported in [32] and [33]. The symbols in Fig. 7(b) represent showninFig.9.TenlaserswereoperatedonthedesiredWDM
measurements with different combinations of lasers, having grid in the C band (193.300–193.525 THz for experiments on
linewidths of approximately 4 kHz, 100 kHz, and 2 MHz. In a 25-GHz grid and 193.35 THz–193.50 THz for a 16.67-GHz
thecontextofourdiscussiononlaserphasenoisetolerance,we grid).Ofthese,9weredistributedfeedback(DFB)lasers,while
also acknowledge the fact that high-speed implementations of a tunable external-cavity laser (ECL) with a 3-dB linewidth
decision directed PLLs may require substantial parallelization of 100 kHz was used for the respective channel under test.
and pipelining which reduces the linewidth tolerance of the The sets of even and odd channels were each generated using
system, as noted in [29]; in this case, feed-forward carrier a separate transmitter of the type shown in Fig. 2. Odd and
recovery schemes can be used as a promising alternative[29], evenchannelswereamplifiedusingerbium-dopedfiberampli-
[30]. One such technique is similar to the ML technique de- fiers (EDFAs) and combined by a symmetric interleaver (IL),
scribedalongwithFig.5(b). 25GHz/50GHzforour25-GHzspacedWDMexperimentsand
16.7 GHz/33.4 GHz for our 16.67-GHz spaced WDM experi-
ments.Inour1022-kmtransmissionexperiment,asimple3-dB
C. Narrow-BandOpticalFiltering
couplerwasusedforWDMcombination,enabledbytheelec-
Figure8characterizesthefilteringtoleranceofoursetup.The tronicwaveformshapingthroughelectricallow-passfilteringof
shaded areas apply to simulations using super-Gaussian filters thedrivesignal.
552 JOURNALOFLIGHTWAVETECHNOLOGY,VOL.28,NO.4,FEBRUARY15,2010
Fig.9. ExperimentalsetupusedforallourWDMexperiments.
Fig.11. Resultsofafilterconcatenationexperimentina10-channel25-GHz
WDM environment (inset). The performance (measured BER expressed in
termsoftheQ-factor)isshownontheleftaxisasafunctionofthenumber
of add/drop nodes passed, and the concatenated filter bandwidth (corrected
comparedto[14])isshownontherightaxis.
At the receiver, the test channel, including a portion of its
neighbors, was selected using a 0.25-nm-bandwidth optical
filter,amplified, and combined with a LOin a coherentdetec-
tionsetupaccordingtoFig.3.
A. FilterConcatenationona25-GHzWDMGrid
Fig.10. Magnitudeandgroupdelayresponseofthe25-GHzinterleaversused
inourexperiments. Figure 11 showsthe results of a filter concatenation experi-
ment[14],representedintermsofthe receivedQ-factor. With
all10channelson(cf.inset),WDMcrosstalkresultsinaBER
AfterWDM-combination,thesignalswerepolarization-mul- floorat (Q-factorof10.38dB).Theobservedperfor-
tiplexed by splitting into two paths and recombining in a mance degradation represents a combination of filter concate-
polarization beamsplitter (PBS), using manual polarization nation, WDM crosstalk, and reduced OSNR as the number of
controllers (PCs) and a decorrelation delay of 20 ns (280 loopround-tripsincreases.After7passesthroughtheadd/drop
symbols). The recirculating loop contained different optical node,the -factorisreducedto9.38dB(BERof ),
elements,dependingontherespectiveexperiment: representing a penalty of 1 dB. Also shown in the figure is
• Inthefilter-concatenationexperiment(SectionV.A,[14]), the concatenated optical filter bandwidth as a function of the
theloopcontainedashort(20-km)spanofstandardsingle- number of nodes passed. For 10 filter passes, the bandwidth
modefiber(SSMF)togetherwithanadd/dropnodecom- shrinks to about 70% of its starting value. Note that the adap-
posed of two 25 GHz/50 GHz interleavers. Fig. 10 gives tiveequalizerinthecoherentreceiveralsocompensateswellfor
magnitudeandgroupdelaycharacteristicsofanindividual asignificantamountofgroupdelayrippleoftheopticalfilters
such interleaver. The 3-dB bandwidth of the full node is (cf. Fig. 10), which adds approximately linearly for each loop
20.6 GHz. round-trip. (‘Approximately’ due to the slight frequency shift
• In the long-haul transmission experiment (Section V.C imposedbytheacousto-opticswitchesforeachlooproundtrip).
[16]) as well as in the ultrahigh spectral efficiency ex-
periment (Section V.D [15]), the loop contained four Then,the10channelsweretransmittedthroughastraightline
80-kmspansofSSMF(totallooplengthof315km)and offour 80-kmspansofSSMF(atotalof315km)withEDFA-
again-equalizingfilter.Thelossofthespansrangedfrom only repeaters and no dispersion-compensating fiber. An add/
16dBto17dB.Amplificationwasprovidedbybackward dropnodewasincludedafterthesecondspan.Theper-channel
RamanpumpingandEDFArepeaterswithapproximately powerlaunchedintoeachspanwas dBm,yieldingaBERof
5-dBnoisefigures.Nodispersion-compensatingfiberwas forthetestchannelatareceivedOSNRof28dB.
used.Inthelong-haultransmissionexperiment,acoupler
B. ElectronicPulse-Shaping:25-GHzWDMWithoutILs
wasplacedaftertheamplifierfollowingthefirstspan,and
was used to sample the signals after transmission over The BER floor due to WDM crosstalk observed in the pre-
thirteenspans(3completeloops km). vious experiment was eliminated by using electrical low-pass
filteringpriortomodulation,whichshapestheopticalspectrum
WINZERetal.:SPECTRALLYEFFICIENTLONG-HAULOPTICALNETWORKING 553
Fig. 14. Transmitted and received WDM spectra on a 25-GHz grid after
1022-km transmission. Upper right: Optimization of Raman gain and
per-channelsignallaunchpower.BERresultsaftertransmissionareshownin
thelowerleftplottogetherwiththereceivedspectrum.Lowerright:Recovered
Fig.12. Effectofelectronicpulseshapingon4-leveldrivesignaleyediagrams
symbolconstellationsbeforeandaftertransmission.
and on 16-QAM optical spectra. Pulse shaping significantly reduces WDM
crosstalk,ascanbeseenfromthe25-GHzspacedneighborsshowningray.
C. Long-HaulTransmissionona25-GHzWDMGrid
We performed transmission of 10 WDM channels on a
25-GHzgridover945km(3fullloops)aswellasover1022km
(3fullloopsplusonespan).Fig.14summarizestheresultsof
the WDM experiment. The figure shows the transmitted and
received WDM spectra after 1022 km (using optical spectrum
analyzers with different resolutions). The upper right portion
of the figure shows the received BER as a function of the
per-channel signal launch power. At 945 km, the optimum
launch power is dBm at a Raman gain of 10 dB. For
Fig.13. BERmeasurementsforcolorless(interleaver-free)WDMcombination 1022-km transmission we increased the Raman gain to 12
andforWDMchannelspacingsof25GHz,20GHz,and17GHz.
dB and found an optimum launch power of dBm. After
1022-kmtransmission,theaverageOSNRofthechannelswas
24dB.TheBERofeachofthetenchannelsisshowntogether
by curtailing its sidelobes. As shown in Fig. 2, simple passive
withthereceivedspectruminthelowerleftportionofFig.14.
Gaussianelectricallow-passfilterswithbandwidthsofapprox-
ThehighestBERis ,belowthecorrectionthreshold
imately11GHzwereusedtothisend,avoidingmoresophisti-
of ofenhancedFECwith7%overhead.Therecovered
catedhigh-speeddigitalpulseshapingthroughdigital-to-analog
signalconstellationsforchannel5areshownatthetransmitter
conversion of programmed raised-cosine waveforms [8], [9].
andafter1022-kmtransmission.
Thefour-leveleyediagramsoftheshapedandunshapedmodu-
latordrivesignalsaswellasofthecorrespondingopticalspectra
D. WDMTransmissionat6.2b/s/HzSpectralEfficiency
areshowninFig.12.
The good suppression of spectral side lobes allowed us to As evident from Fig. 13, going below 25-GHz channel
combineWDMchannelsona25-GHzWDMgridwithoutthe spacing requires additional optical interleaver filtering at the
useofinterleavers,usingasimple3-dBcoupler.Fig.13shows transmitter to avoid significant WDM crosstalk penalties. In
BER measurements for this case. The triangles represent the ordertoachieve16.67-GHzchannelspacingwithlowpenalties,
reference single-channel measurement of PDM 16-QAM, re- we therefore used a custom-designed silica-on-silicon, 0.80%
vealing a back-to-back required OSNR of 20.8 dB to achieve normalized-index-step planar-lightwave circuit with highly
aBERof ,whichisabout1.2dBbetterthanourfirst sampled arrayed-waveguide gratings (AWGs) in a configura-
results shown in Fig. 6, and is off the theoretical limit (solid tion similar to [34] and [35]. It consists of two AWGs with
line) by 3.8 dB. In 25-GHz WDM operation (circles), WDM a free-spectral range of 33.33 GHz and 14 grating arms each
crosstalkfromthetwonearestneighborsimposesanadditional connected by 7 equal-length waveguides. The magnitude and
penalty of 1 dB. The penalty rises rapidly for WDM channel group-delayresponsesoftheinterleaverpassbands,witha3-dB
spacingsbelow20GHz(seecurvesinFig.13for20-GHzand bandwidthof13GHzareshowninFig.15.
17-GHzspacing).Therecovered16-QAMsignalconstellations Fig. 16 shows back-to-back BER measurements. The tri-
ofacentralWDMchannelat25-GHzspacingand21-dBOSNR angles represent the measurement of a single-channel PDM
(BERof )arealsoshown. 16-QAM signal without using the 16.67-GHz/33.33-GHz
554 JOURNALOFLIGHTWAVETECHNOLOGY,VOL.28,NO.4,FEBRUARY15,2010
Fig.15. Magnitudeand group delayresponse ofthe16.67-GHz interleaver
usedinourexperiments.
Fig.17. TransmittedandreceivedWDMspectraona16.67-GHzgridafter
630-kmtransmission.BERresultsaftertransmissionareshowninthelower
leftplottogetherwiththereceivedspectrum.Lowerright:Recoveredsymbol
constellationsaftertransmission.
16-QAM signals at 112 Gb/s. We have described transmitter
and receiver setups as well as the underlying detection algo-
rithms in detail and analyzed their performance using both
simulated and measured waveforms. Our receiver requires an
ADC resolution of 5 bits and a frequency difference between
signal and LO lasers of below 20 MHz. The aggregate
Fig. 16. BER measurements of 16-QAM for 16.67-GHz spaced WDM.
linewidth of the two lasers must stay below several hundred
Solidcurve:Theory;Triangles:Singlechannel,nointerleaver;Circles:Single
channel,16.67-GHzinterleaver;Squares:WDMona16.67-GHzgrid. kHz in order to avoid noticeable OSNR penalties. The entire
setupyieldsaback-to-backrequiredOSNRof20.2dB,which
is3.2dBoffthetheoreticallimit.
interleaver, revealing a back-to-back required OSNR
Ina25-GHz-spaced10-channelWDMenvironment(4.1b/s/
of 20.2 dB, another 0.6-dB improvement
Hz), we measured a 1-dB -factor penalty after seven passes
compared to the results of Section V.B and only 3.2 dB off
throughanadd/dropmultiplexer,andtransmittedthesignalover
the theoretical limit, solid line. When passing through the
1022 km of uncompensated, hybrid EDFA-Raman amplified
interleaver(circles), a penalty of 1.3 dB is incurreddue to the
80-kmspansofSSMF.Electricalwaveformshapingallowedus
tightopticalfiltering,inlinewiththeresultsofFig.8.Crosstalk
tocombinetheWDMchannelsusingasimple3-dBcouplerand
from the 33.3-GHz spaced neighbors within the same set of
noopticalinterleaverfilters.
WDM channels (i.e., for operation with even channels only
Ona16.67-GHzWDMgrid,weachieved10-channeltrans-
or with odd channels only) was measured to be negligible. In
mission over 630 km using the same line system as for the
16.7-GHz WDM operation (squares), crosstalk from the two
1022-km experiment. This yields a high spectral efficiency of
nearest neighbors imposes an additional penalty of 1.5 dB.
6.2b/s/Hz,with1-Tb/sofinformationfittingwithin1.2nmof
Fig.16alsoshowstherecovered16-QAMsignalconstellations
opticalbandwidth.Aspectralefficiency distanceproductof
for both polarizations of a single unfiltered channel at high
3906b/s/Hz kmwasobtained.
( 35 dB) OSNR as well as for a central WDM channel at
23-dBOSNR(BERof ). ACKNOWLEDGMENT
After630-kmtransmission(2loopround-trips;10-dBRaman
TheauthorsthankR.W.Tkachforvaluablediscussions;V.
gain; dBm per-channel signal launch power), the average
FaraciandG.Rodetforproviding25-GHzinterleavers;andM.
OSNR of the channels was 24 dB. Transmitted and received
Cappuzzo,E.Y.Chen,L.T.Gomez,F.Klemens,andR.Keller
spectraareshowninFig.17(usingopticalspectrumanalyzers
for16.67-GHzsilicawaveguideinterleaverfabrication.
withdifferentresolutions).Thereceivedsignalconstellationsof
acentralchannelarealsoshown,togetherwiththeBERofeach
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“Transmission of 32-Tb/s Capacity Over 580km Using RZ-Shaped Vienna,Austria,in1998.Hisacademicwork,largely
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tems based on IQ-transmitters and homodyne receivers with digital oftheInstituteofElectricalandElectronicEngineers(IEEE),andformerAs-
phaseestimation,”inProc.OFC’06,2006,PaperNWA4. sociateEditorforIEEEPhotonicsTechnologyLetters(2000–2009).
556 JOURNALOFLIGHTWAVETECHNOLOGY,VOL.28,NO.4,FEBRUARY15,2010
Christopher R. Doerr (F’06) received the B.S. Labs, Alcatel-Lucent, Holmdel, NJ. He has authored and coauthored more
degree nautical/astronautical engineering and the than30journalandconferencepapers.Hisresearchinterestsareinthebroad
B.S., M.S., and Ph.D. degrees (1995) in electrical areaofcommunicationtheory.Topicsincludesynchronization,channelesti-
engineering,allfromtheMassachusettsInstituteof mation, equalization, coding and reduced complexity detection schemes for
Technology(MIT),Cambridge.HeattendedMITon multi-antennasystems.
anAirForceROTCscholarshipandearnedhispilot
wingsatWilliamsAFB,Arizona,in1991.HisPh.D.
thesis,onconstructingafiber-opticgyroscopewith
noise below the quantum limit, was supervised by LawrenceL.Buhl(Larry)receivedaB.Sc.inelec-
Prof.H.Haus.
tronicengineeringfromMonmouthUniversity,Long
SincejoiningBellLabsin1995,Doerr’sresearch
Branch,NJ,in1984.
hasfocusedonintegrateddevicesforopticalcommunication.Hewaspromoted He first served in the US Army, Signal Corp,
toDistinguishedMemberofTechnicalStaffin2000,
Fort Monmouth, NJ, as an instructor, teaching
Dr.DoerrreceivedtheOSAEngineeringExcellenceAwardin2002,became Microwave theory and Radar systems. He joined
anIEEEFellowin2006andanOSAFellowin2009,andreceivedintheIEEE
BellLaboratoriesin1972,beginningearlyresearch
William Streifer Awardin 2009. He was Editor-in-Chiefof IEEE Photonics
work in metal vapor laser spectroscopy. In 1978,
TechnologyLettersfrom2006–2008andiscurrentlyanAssociateEditorfor
hebeganpioneeringand developed variousguided
theIEEE/OSAJOURNALOFLIGHTWAVETECHNOLOGY. wave optical devices in LiNbO and later, III-V
materials.Since2000,hehasbeeninvestigating10G
and40Gsystemsanddevicesandrecentlyassumedresponsibilityforoptical
andhighspeeddevicepackaging.
Maurizio Magarini was born in Milano, Italy, in
1969. He received the M.S. and Ph.D. degrees in
electronic engineering from the Politecnico di Mi-
lano,Milano,Italy,in1994and1999,respectively.
In 1994, he was granted the TELECOM Italia
scholarship award for his Master thesis. From
1999 to 2001 he was a Research Associate in the
Dipartimento di Elettronica e Informazione at the
Politecnico di Milano where, since 2001, he has
been anAssistant Professor. From August2008 to
January 2009 he spent a sabbatical leave at Bell