Nonlinear magnetoinductive transmission lines

2.9 MB·

by  Nikos Lazarides

NONLINEAR MAGNETOINDUCTIVE TRANSMISSION LINES
NIKOS LAZARIDES†, VASSILIS PALTOGLOU‡ and G. P. TSIRONIS⋆
Department of Physics, University of Crete, and FORTH
P.O. Box 2208, 710 03, Heraklion, Crete, Greece
†[email protected]
‡[email protected]
⋆
[email protected]
Power transmission in one-dimensional nonlinear magnetic metamaterials driven at one end is
investigatednumerically andanalytically inawidefrequencyrange. Thenonlinearmagneticmeta-
1
materials are composed of varactor-loaded split-ring resonators which are coupled magnetically
1
through their mutual inductances, forming thus a magnetoiductive transmisison line. In the lin-
0
ear limit, significant power transmission along the array only appears for frequencies inside the
2
linear magnetoinductive wave band. We present analytical, closed form solutions for the magne-

n toinductive waves transmitting the power in this regime, and their discrete frequency dispesion.
a When nonlinearity is important, more frequency bands with singificant power transmission along
J the array may appear. In the equivalent circuit picture, the nonlinear magnetoiductive transmis-

3 sion line driven at one end by a relatively weak electromotive force, can be modeled by coupled
1 resistive-inductive-capacitive(RLC)circuitswithvoltage-dependentcapacitance. Extendednumer-
ical simulations reveal that power transmisison along the array is also possible in other than the

] linearfrequencybands,whicharelocatedclosetothenonlinearresonancesofasinglenonlinearRLC
h
circuit. Moreover, the effectiveness of power transmission for driving frequencies in the nonlinear
p
bands is comparable to that in the linear band. Power transmission in the nonlinear bands occurs
-
s throughthelinear modesofthesystem,andit isclosely related totheinstability of amodethatis
s localized at thedriven site.
a
l
c PACSnumbers:
. Keywords: nonlinear split-ring resonators, magnetic metamaterials, driven linear waves, nonlinear power
s
transmission
c
i
s
y I. INTRODUCTION materials. Magnetism at such high frequencies is partic-
h
ularly important for the implementation of devices such
p
as tunable lenses, adaptive mirrors, etc., since there are
[ The metamaterial concept usually refers to a periodic
only a few natural materials that respond magnetically
arrangementof artificially structured elements, designed
1 above microwaves. Moreover, magnetism in those natu-
v to achieveadvantageousand/orunusualelectromagnetic
ralmaterialsisusuallyweakandwithinnarrowfrequency
9 propertiescomparedtothoseofnaturallyoccuringmate-
bands, limiting thus their use in possible Terahertz de-
1 rials [42]. For example, the first metamaterial ever real-
vices. The most common realization of such a magnetic
5 ized [40], exhibited negativerefractionindex in a narrow
2 metamaterial iscomposedofperiodicallyarrangedSRRs
frequency band around 5 Gigahertz. The negative re-
. in a one- ortwo-dimentionallattices. In its simplest ver-
1 fraction index property requires that both the dielectric
sion an SRR is just a highly conducting [5, 13, 24] or
0 permittivity and the magnetic permeability are simul-
1 superconducting [8] ring with a slit. The SRR structure
taneously negative. The most widely used elements for
1 wasoriginallyproposedbyPendry[30],whotheoretically
constructinganegativerefractiveindexmetamaterialare
: predicted that a periodic SRR-based structure could ex-
v the electrically small resonant’particles’called split-ring
hibit a negative magnetic permeability frequency band
i resonators (SRRs)andmetallicwires. Whilethearrayof
X when irradiated by an alternating electromagnetic field
wiresgivesa negativepermittivity below the plasma fre-
r ofappropriatepolarization. Pendryalsorealizedthatthe
a quency, the arrayof the SRRs gives a negative magnetic
SRR structure has considerable potential for enchancing
permeability above their resonance frequency. Thus, the
nonlinear phenomena. When the SRR dimensions are
refraction index can be negative in a narrow frequency
much smaller than the wavelength of the incident field,
band. Sincetheirdiscovery,metamaterialshavebeenthe
the SRR canbe regardedasanelectric circuitconsisting
target of intensive study. Although the developement of
ofinductive,capacitiveandohmicelementsconnectedin
metamaterials at microwave frequencies has progressed
series. The ring forms the inductance L, while the slit
tothepointwherescientistsandengineersarenowpursu-
can be considered as a capacitor of capacitance C. The
ing applications, researchon metamaterials that operate
ohmic element R may represent all possible losses of the
athigherfrequenciesis stillinanearlystage[25,26,43].
structure.
Thereisawidesubclassofmetamaterials,thatexhibit
significant magnetic properties and even negative mag- Thenextstepwasnaturallytobuiltnonlinearityinthe
netic permeability at Terahertz and optical frequencies SRR structure. Nonlinearity implies real-time tunability
[13, 25, 51], even though they are made of non-magnetic and multistability that is certainly a desired property of
2
possiblefuture devices. TheSRRs canbecomenonlinear toinductive wave modes, slightly modified by the pres-
either by the insertion of a strongly nonlinear dielectric enceofnonlinearity. However,forfrequenciesinthepass-
[9] or a nonlinear electronic component (i.e., a varactor) bandsresultingfromnonlinearity,the poweris transmit-
[33,34,50]intotheirslits. Bothwayswouldleadtoeffec- ted along the array from nonlinear, breather-like exci-
tivelyfielddependentmagneticpermeabilityforanarray tations. In the next Section we derive the model equa-
of nonlinear SRRs. Recently, the dynamic tunability by tions for a varactor-loaded SRR-based magnetic meta-
anexternalfieldfora two-dimensionalarrayofvaractor- material, where the nonlinearity has the form a polyno-
loadedSRRswasdemonstratedexperimentally[37]. The mial expansion on the main variable, like that investi-
SRRs in such an array are coupled through magnetic gated in [50]. We also present bifurcation diagrams for
and/orelectricdipole-dipoleinteractions,whosestrength the shortestpossibleend-drivenarray,i.e.,anarraywith
dependsontherelativeorientationoftheSRRsandtheir only two varactor-loaded SRRs, from which only one is
slits in the array [6, 10, 31, 45]. directly driven. In Section III we obtain analytical solu-
tions for the linear magnetoinductive waves of the end-
In the equivalent circuit picture, the dynamics of a
driven,finite SRRarraywhenlossesareneglected,along
periodic nonlinear SRR array irradiated by an electro-
with their discrete dispersion relation. In Section IV we
magnetic field, can be described by a set of coupled and
presentnumericalresultsforpowertransmissionanddis-
driven ordinary differential equations with on-site non-
cuss their dependence on the model parameters for a
linearity. The discreteness, which is inherent in those
short, end-driven varactor-loaded SRR array with and
SRR-basedmagneticmetamaterials,alongwiththe non-
without an absorbing boundary at the non-driven end.
linearity and the weak coupling between their elements,
We finish with concluding remarks in Section V.
allow for the generation of nonlinear excitations in the
form of intrinsic localized modes or discrete breathers
[41]. These nonlinear modes appear generically in dis-
II. VARACTOR-LOADED SRR ARRAY MODEL
crete and nonlinear extended systems. Recent theoreti-
calworkinone-andtwo-dimensionalnonlinearSRRlat-
ticeshavedemonstratedtheexistenceandthestabilityof Consider a ring-shaped or a squared split-ring res-
discretebreathersbothinenergy-conservinganddissipa- onator with a hyperabrupt tuning varactor mounted
tive systems [3, 4, 16, 18]. It has also demonstratedthat onto its slit. The varactor type could be selected
breathers may be formed spontaneously through modu- from a large variety of available varactors, according
lational instability in binary nonlinear magnetic meta- to the needs of the experiment. In recent experimen-
materials[20,22,29]. Moreover,domain-wallexcitations tal works on varactor-loaded SRR-based metamateri-
[35] and envelope solitons [1, 2, 14, 48] may as well be als [11, 15, 32, 50], the varactor selected was a Sky-
excited in those systems, which seem to be stable even works SMV1231-079, whose voltage-dependent capaci-
in the presence of noise [49]. tanceC(UD)(UD isthevoltageacrossthediode)isgiven
by
The SRR-based magnetic metamaterials support a
new kind of electromagnetic waves, the magnetoinduc- U −M
D
tive waves, which exhibit phonon-like dispersion curves, C(UD)=C0 1 , (1)
− U
andthey cantransferenergyalongthe array[39,46,47]. (cid:18) p(cid:19)
It is thus possible to fabricate a contact-free data and where C0 is the DC rest capacitance, Up is the intrinsic
power transfer device, a magnetoinductive transmission potential,andM isaparameter. Thevaluesofthosepa-
line, which make use of the unique properties of the rametersareprovidedbythemanufacturerSPICEmodel
magnetic metamaterial structure, and may function as for that varactor to be
a frequency-selective communication channel for devices
M =0.8, C =2.2 pF, U =1.5 V. (2)
via their magneto-inductive wave modes [44]. In the 0 p
presentworkweinvestigatethepowertransmissionalong From Eq. (1) we can determine the voltage-dependence
a one-dimensional varactor-loaded SRR-based magnetic of the normalized charge
metamaterial, in the two possible geometries, which is
Q
driven at one end by a sinusoidal power sourse. In the q = D , (3)
C U
linearlimit,energytranferalongthearrayoccursonlyfor 0 p
driving frequencies in the linear magnetoinductive wave with Q being the charge in the diode, as a function of
D
band. However,thenonlinearitycouldgeneratemorefre- the normalized diode voltage u=U /U as
D p
quency bands where efficient power transmission along
1
the array is possible. Power transmission in chains of q = 1 (1 u)−M+1 . (4)
coupled anharmonic oscillatorsfor driving frequencies in M +1 − −
− h i
the band gap of the linear spectrum has been recently Assuming that U < U , the earlier equation can be
D p
investigated,andthateffectis referredto asself-induced solved for the diode voltage as a function of the charge,
transparency [28] or supratransmission [7]. For frequen- i.e., u=u(q)
cies inside the linear magnetoinductive wave band the
1
power is transmitted along the array from the magne- u=1 [1 q( M +1)](−M+1) . (5)
− − −
3
When x q( M+1) <<1,thevoltageuintheearlier 5
| |≡| − |
equation can be expanded in a Taylor series inpowers of
q around zero. Thus, neglecting term of order (q4) or 4
O
higher, we get
3
u=q+αq2+βq3, (6)
0
i
where, by using Eq. (2), 2
M 1
1
α= = 0.4, β = M(2M 1)=+0.08. (7)
− 2 − 6 −
The expression Eq. (6) can then be used in the volt- 0
0 1 2
age equation, obtained from Kirchhoff’s voltage law, Ω
whichrepresentsthevaractor-loadedSRRasaneffective
resistive-inductive-capacitive(RLC)circuitwithvoltage- FIG.1: Thecurrentamplitudei0asafunctionofthenormal-
dependent capacitance ized frequency Ω for a varactor-loaded SRR modeled by Eq.
(13). Red curve: frequency increases; black curve: frequency
d2Q dQ
D D decreases.
L +R +U u= (t), (8)
dt2 dt p E
where
driven by a sufficiently strong external field. A typical
Q Q 2 Q 3 current amplitude i0 - frequency Ω curve for the equiva-
U u U =U D +α D +β D , lentcircuitmodelEq. (13),likethatshowninFig. 1,ex-
p D p
≡ "C0Up (cid:18)C0Up(cid:19) (cid:18)C0Up(cid:19) # hibitsallthosecharacteristics. Noticethehysteresisloop
(9) at normalized frequency Ω around 1.1, close to the reso-
, (t) is the induced electromotive force of amplitude 0 nancefrequencyofthelinearsystem,andtheverystrong
E E
and frequency ω resulting from an applied electromag- resonance at its second harmonic, i.e., at Ω 2, where
netic field with the same frequency, of the form a smaller hysteresis loop also appears. Furth≃ermore, we
observe a subharmonic resonance at Ω 0.48, as well as
E(t)=E0 sin(ωt), (10) weaker resonances at Ω 0.63, 0.33 an≃d Ω 0.25. For
∼ ∼
frequency intervals lying between the boundaries of the
where L is the inductance of the ring, R is the sum of
hysteresis loops there are two different states of the os-
the Ohmic resistance of the SRR ring and the series re-
cillator, with low and high current amplitude, which are
sistance of the diode, and t the temporal variable. The
simultaneously stable.
eigenfrequency ω of the circuit described by Eq. (8) is
0
given, in the linear limit without losses and driving, by
H D
1 E
ω = . (11) w
0 √LC0 h l
d
That approximate model of the varactor-loaded SRR is
H
foundtobe adequateforvaractorvoltagesnotexceeding
E
0.5V (u 1/3innormalizedunits),whenthedissipative
≤
current of the varactor can be neglected [50].
Using Eq. (3) and the relations
FIG. 2: Schematic of an SRR array in the planar geometry
ω (upper)and in the axial geometry (lower).
0
τ =tω , Ω= , ε = E , (12)
0 0
ω U
0 p
Consider a one-dimensional periodic array of identical
Eq. (8) can be written in the normalized form varactor-loadedSRRs,whicharecoupledmagneticallyto
their first neighbors through their mutual inductances.
d2q dq
+q+αq2+βq3+γ The array can be formed in two possible geometries, ac-
dτ2 dτ cordingtotherelativeorientationofitselementaryunits
=ε0sin(Ωτ), (13) (i.e., the SRRs) in the array. As can be seen in Fig. 2,
the SRRs can be arranged either in the planar geome-
where
try, where all of them are lying on the same plane, or in
the axial geometry, where the axes of all the SRRs are
γ =R C /L (14)
0
lying on the same line. The orientation of the incident
is the normalized loss coeffipcient. The varactor-loaded electromagneticfield,whichisalsoshowninthefigure,is
SRR is a nonlinear oscillator, which exhibits multista- suchthatthe magnetic componentcanexcite anelectro-
bility, hystereticeffects,andsecondaryresonances,when motiveforceineachSRR. Belowweconsiderarraysthat
4
aredrivenonlyatoneend(letus saytheleftend). Thus 20 (a)
only one SRR receives energy directly from the driver. 15
Hduoewetovetrh,ethienteenrearcgtyiocnanbebtweeterannistmsietlteemdeanlotsn.gItnhceluadrrinagy Etot10
thecouplingofeachSRRtoitsnearestneighbors,weget 5
for the charge Q of the n th varactor the dynamic
D,n
− 0
equations
1
(b)
dI dI dI
n n−1 n+1
L +RIn+Upun+M +M = δn,1(,15) 2
dt dt dt E f
0.5
,
1
where n = 1,...,N, with N being the total number of f
SRRs in the array, I (t)=dQ /dt is the currentflow-
n D,n 0
ing in the n th SRR, U u is the approximate voltage
p n
across the n−th varactor, and M (not to be confused 1 (c)
−
with the varactor parameter) is the mutual inductance
2
betweentwoneighboringSRRs. Usingtherelationsfrom f
0.5
,
Eq. (12), thecouplednonlinearEqs. (15)canbewritten 1
f
in normalized form as
0
d2 1 2
dτ2 (λqn−1+qn+λqn+1)+qn+αqn2 +βqn3 = Ω
dq
n
γ +ε sin(Ωτ)δ , (16)
− dτ 0 n,1 FIG.3: (a) Energy Etot bifurcation diagram forvaryingfre-
quency Ω for an asymetrically driven varactor-loaded SRR
whereλ=M/Listhecouplingcoefficientbetweenneigh- dimer with γ = 0.001, ε0 = 0.3, and λ = +0.01. (b) Energy
boring SRRs. The function δn,1 is unity for n=1, while fractions f1 and f2 of thefirst (driven) and the second SRR,
itiszeroforanyothern. Thecouplingparameterλmay respectively,ofthedimerinthelow energystate. (c)Energy
assume bothpositive and negativevalues,corresponding fractions f1 and f2 of thefirst (driven) and the second SRR,
toplanarandaxialgeometryofthevaractor-loadedSRR respectively, of thedimer in thehigh energy state.
array,respectively.
For a particular array, the values of the coupling and
loss coefficients λ and γ, respectively, can be estimated (19)thatRSRR 0.13Ω. Thediode seriesresistanceRs
≃
from its geometrical and material parameters. Specifi- of the varactor used in Ref. [50] is around 2 Ω; that
≃
cally, the mutual inductance M, the ring inductance L, is, most of the Ohmic resistance of the varactor-loaded
and the Ohmic resistance RSRR of the ring can be cal- SRR,R=RSRR+Rs =0.13+2.00=2.13Ωcomesfrom
culated from the following expressions the varactor. By substituting R 2.13 Ω in the second
≃
of Eq. (14), along with L=14 nH and C =2.2 pF, we
0
πa2a2 get that γ = 0.027. The resonance frequency of a single
M = µ , (17)
0 4D3 varactor-loaded SRR is f0 (2π√LC0)−1 0.9 GHz.
≃ ≃
16a In the numerical calculations below we used the esti-
L = µ a ln 1.75 , (18)
0 mated value of λ that indicates weakly coupled SRRs.
h −
(cid:20) (cid:18) (cid:19) (cid:21) However, we use a smaller value of γ, which could be
2a
R = , (19) achieved in practice by using a metalic wire with higher
SRR
σhδ
conductivity and an ultra-low resistance varactor with
where µ is the magnetic permeability in vacuum, a is similar capacitance-voltagerelation.
0
the average radius of each SRR, h the diameter of the The shortest end-driven varactor-loadedSRR array is
wire of each SRR, D is the center to center distance be- thatcomprisedoftwoelements. Sinceonlyoneofthemis
tweenneighboringSRRs,andσ andδ istheconductivity drivenbytheexternalfield,werefertothissystemasthe
and skin depth of the SRR wire. Eqs. (17)–(19) refer to asymmetrically driven nonlinear dimer. The increase of
circular SRRs with circular cross-section. However,they degrees of freedom leads to interesting and complicated
can also be used for circular SRRs with square cross- behavior, that is marked by the appearance of chaos. In
section by defining an equivalent diameter of the SRR Fig. 3,thebifurcationdiagramofthetotalenergyE of
tot
wire h = 4dw/π, with d and w being the depth and the dimer with varyingnormalizedfrequency Ω is shown
width, respectively, of the SRR wire. For a 3.7 mm, for positive coupling λ. For negative coupling λ with
p ≃
D 11 mm, h 0.5 mm, we get from Eqs. (17) and equal strength we get very similar results that share a
≃ ≃
(18), respectively, that M 0.14 nH and L 14 nH, number of characteristic features. First of all, multista-
and consequently λ 0.01≃. For δ 2.2 ≃10−3 mm bility is again present, for normalized frequencies in an
and σ 0.5 106 (Ω≃cm)−1 (for an SR≃R mad×e of copper interval around Ω 0.9. There, two simultaneously sta-
≃ × ∼
wire at 1 GHz at room temperature) we get from Eq. bleenergystates,withhighandlowenergy,co-exist,asit
∼
5
isshowninFig. 3a. Theenergydifferencebetweenthose the structure, i.e.,
states increases condiderably with increasing frequency.
Moreover,low energy state becomes chaotic for frequen- q0 =qN+1 =0. (22)
cies around Ω 1.9. It is also interesting to see how the
∼ By substituting q =Q sin(Ωτ) into Eq. (20), the sta-
total energy is divided between the two elements. The n n
tionary equations can be written as
energy fractions f = E /E and f = E /E , with
1 1 tot 2 2 tot
E and E being the energies of the first (driven) and
1 2 sQ +Q +sQn+1=κ, (23)
n−1 n
second SRR of the dimer, are shown both for the high
and the low energy states in Fig. 3b and 3c as a func- where
tion of the normalized frequency Ω. When the dimer is
in the low energy state, most of the energy is concen- λΩ2 ε0
s= , κ= δ , (24)
trated in driven SRR whose energy fraction is close to −1 Ω2 1 Ω2 n,1
− −
unity (correspondingly, the energy fraction of the other
or, in matrix form
SRR is close to zero). However, for frequencies around
the resonances, energy can be transfered easily from the Q=κSˆ−1E , (25)
1
one SRR to the other, and then the energy fractions of
the SRRs in the dimer attain comparable values. In the where Q and E are N dimensional vectors with com-
1
chaotic regions, the energy is tranfered irregularly from ponets Q and δ , resp−ectively, and Sˆ−1 is the inverse
i i,1
one SRR to the other and the energy fractions span the of the N N coupling matrix Sˆ. The latter is a real,
whole interval between zero and one. In the high energy symmetric×tridiagonalmatrixthathasdiagonalelements
state (Fig. 3c), the energy fractions are almost equal for
equal to unity, while all the other non-zero elements are
frequencies above the linear resonance frequency, Ω 1. equal to s. The need to find the inverse of tridiagonal
∼
Belowthatfrequency,theenergyiseitherconcentratedin matrices like Sˆ arises in many scientific and engineering
the driven SRR (for frequencies far from resonances), or applications. Recently, Huang and McColl [12] related
the values of the energy fractions are comparable (close the inversion of a general trigiagonal matrix to second
to resonances. order linear recurrences,and they provided a set of very
simpleanalyticalformulaefortheelementsoftheinverse
matrix. Thoseformulaeleadimmediatelytoclosedforms
III. DRIVEN LINEAR SOLUTIONS for certain trigiagonal matrices, like Sˆ [21]. The compo-
nents of the Q vector can be written as
For a weakly driven array, the nonlinear terms which
are proportional to α and β are not so important and Q =κ Sˆ−1 , (26)
i
they can be neglected. Then, the system of Eqs. (16) i1
(cid:16) (cid:17)
reduces to a linear one
where Sˆ−1 is the (i1) element of the inverse cou-
d2 (λq q +λq )+q pling m(cid:16)atrix(cid:17)iSˆ1−1, whose e−xplicit form is given in [21].
dτ2 n−1− n n+1 n Then, the solution of the linear system Eq. (20) and
dq
n (22) for any driving frequency and finite N is given by
= γ +ε sin(Ωτ)δ . (20)
0 n,1
− dτ
sin[(N n+1)θ′]
The dispersion relation of the linear system can be ob- qn(τ)=κµ sin[(N−+1)θ′] sin(Ωτ),
tained by the substitution q = Acos(κn Ωτ) in the
n
− 1
absence of losses and applied field, i.e., for ε0 = 0 and θ′ =cos−1 , (27)
γ =0, which gives the linear dispersion of magnetoiduc- 2s
(cid:18) | |(cid:19)
tive waves
for s>+1/2 and s< 1/2, and
−
1
Ω = , (21)
κ sinh[(N n+1)θ]
1+2λ cos(κ) q (τ)=κµ − sin(Ωτ),
n
sinh[(N +1)θ]
p
where κ is the (normalized) wavevector ( π κ π). 1+ 1 (2s)2
Eq. (21) defines a frequency band of wi−dth≤∆Ω ≤ 2λ θ =ln − , (28)
thatisboundedbyaminimumandamaximumfrequ≃ency p2|s|
Ωmin =1/√1+2λandΩmax =1/√1 2λ,respectively. for 1/2<s<+1/2, where
However,itis alsopossibleto calcula−te the dispersion, −
expressedas a series of resonantfrequencies, and the ex- 1 s n−1
µ= | | . (29)
act form of the linear magnetoinductive modes for the
s − s
finite system in the absence of losses (γ = 0). For finite | |(cid:18) (cid:19)
systems, Eq. (20) should be implemented with free-end Note that the Q s are uniquely determined by the pa-
i
boundary conditions, to account for the termination of rameters of the system.
6
20
(a)
n
10 150 Q 10
0
8 100
40
(b)
50
6 Qn 0
n
0
4 -40
-50 40
(c)
2
-100 Qn 0
00.99 1 1.01 -150 -40
Ω 40
(d)
n
Q 0
FIG.4: DensityplotsoftheQn’sinastationarystate,onthe
sitenumbern-normalizedfrequencyΩplanecalculatedfrom -40
the analytical expressions Eqs. (27) and (28) for λ = +0.01 20
andε0 =0.3. Theverticalwhitebandsindicatethepositions
n 0
of the resonances. Q
(e)
-20
From the analytical solution in the linear magentoin- 2 4 6 8 10
n
ductive wave band, Eq. (27), which corresponds to ei-
ther s > +1/2 or s < 1/2, we can infer the resonant
−
frequenciesfromzeroingthedenominator,i.e.,bysetting FIG. 5: Typical profiles calculated analytically from Eqs.
sin[(N +1)θ′] = 0. Then we see that the solution has a (27) and (28) for λ = +0.01, ε0 = 0.3, and five different
frequencies: (a) Ω=0.989; (b) Ω=0.995;(c) Ω=1.000; (d)
resonance for
Ω = 1.005; (e) Ω = 1.011. The frequencies in (a) and (e) lie
1 outside thelinear magnetoinductivewave band.
s s = , (30)
m
≡ 2cos mπ
(N+1)
h i see in Fig. 4 that the solutions close to the lower bound
where m is an integer (m = 1,...,N). The resonant fre- (Ω 0.99)ofthe linearmagnetoinductivewaveband
min
qquencies are obtained by solving the first of Eq. (24) exhibit≃very high amplitudes. For negative coupling co-
withrespecttoΩ,andsubstitutingtheresonantvaluesof efficientwithequalstrengththehighamplitudesolutions
sdi≡spesrmsifornomreElaqti.o(n30o)f.lTinheeanr wmeaggentettohiendduiscctrievteewdyanveasmiinc alipnpeaerarmcalogsneettoointdhuectuipvepewrabvoeubnadnd(.Ωmax ≃ 1.01) of the
the varactor-loaded SRR array, the discrete analogue of InFig. 5theprofilesoftheanalyticalsolutionsforfive
Eq. (21), as different frequencies, both inside and outside the linear
magnetoinductive wave band are shown for positive λ.
1
Ω = , (31) The frequencies in Figs. 5a and 5e lie outside the lin-
m
1 2λ cos mπ ear band, so that the Qn’s decrease exponentially with
− N+1 increasing site number n.
r
(cid:16) (cid:17)
wheremisthemodenumber(m=1,...,N). InFig. 4,a
densityplotoftheQ ’sonthen Ωplane,calculatedan-
n IV. POWER TRANSMISSION ALONG THE
−
alytically from Eqs. (27) and (28), is shown for positive ARRAY.
coupling coefficient λ. In the frequency interval shown,
theQ ’shaveratherlargevaluesbecause,duetothenar-
n In the following we focus on the power transmission
row band, all frequencies are very close to a resonance.
along a varactor-loaded SRR array with N = 10 ele-
Moreover, the analytical solutions presented above have
ments, which is driven at the left end (n = 1). The
been obtained in the absence of losses, so that there is
average power dissipated in the n th SRR is defined as
nothing to prevent the Q ’s from increasing indefinitely −
n
closetoaresonance(andtogotoinfinityexactlyatreso- P =R<I2(τ)> , (32)
n n τ0
nance). The white verticalregionsin these figures corre-
spond to frequency intervals around a resonance, where where <> denotes time-average over τ time-units.
τ0 0
the Q ’s have very large values. Thus, those regions in- The power density P can be normalized to P = I2R,
n n 0 0
dicate the frequencies of the resonances. We can also where I = ω C U and R the value of resistance that
0 0 0 p
7
results from the value of the dissipation coefficient used
in the simulations. For γ =0.001 we have that R 0.06
≃
andP0 28µW. Itis convenientto expressPn indBm, 10 0
≃
according to the relation
8 -50
P (in W)
P (dBm)=20log <i2 > 0
n (cid:18) n τ0 1 mW (cid:19) n 6 -100
≃4.34ln(0.028<i2n >τ0), (33) -150
4
where i is the normalized current in the n th SRR.
n − -200
2
-250
10 50 0 0.5 1 1.5 2 -300
f (GHz)
0
8
-50
n 6 FIG. 7: Average power density plot (in dBm) on the n−f
-100 plane, for λ = +0.01, γ = 0.001, α = −0.4, β = 0.08, and
4 -150 ǫ0 =0.3.
-200
2
-250 thelosscoefficienthasbeenincreasesbyanorderofmag-
nitude (from 0.001 to 0.01. In the density plots shown
0 0.5 1 1.5 2 -300 above,significant power transmissionis indicated by red
f (GHz)
color. In both Figs. 7 and 8, nonlinear power trans-
mission occurs at f 0.3 GHz, f 0.45 GHz, and
∼ ∼
f 1.8 GHz. The latter nonlinear band, around the
FIG. 6: Average power density plot (in dBm) on the n−f ∼
secondharmonicofthe resonancefrequencyofthe linear
plane, for λ = +0.01, γ = 0.001, and ǫ0 = 0.3, in the linear
system, seems to be the wider of all.
regime (α=0, β=0).
The system of Eqs. (16) is integrated with a 4th
order Runge-Kutta algorithm with fixed time-stepping,
and boundary conditions q = q = 0 to account for 10 0
0 N+1
the termination of the structure. The initial conditions -50
8
were set to zero. The time allowed for the elimination
-100
of transients in each value of the frequency is 2000 T (
0
n 6 -150
T = 2π/ω ), while the time interval τ over which the
0 0 0
averagepoweriscalculatedistypically1000T . Inorder -200
0
4
toconverttimeintervalstophysicalunitsonehastomul- -250
tiplybytheinverseoftheresonancefrequencyf−1,which
0 -300
2
gives τ 2.2 µs. In the linear regime, i.e., α = 0 and
0
≃ -350
β =0,theaveragepowerP(dBm)densityplotonthesite
number n - frequency f plane shown in Fig. 6 exhibits 0 0.5 1 1.5 2 -400
the expected behavior; that appreciable power transmis- f (GHz)
sion only occurs for frequencies in the linear band, i.e.,
around f = 0.9 GHz. However, the activation of the
nonlinear terms reveals that even a relatively low driv- FIG. 8: Average power density plot (in dBm) on the n−f
ing amplitude ε = 0.3 is sufficient for nonlinear power plane, for λ = +0.01, γ = 0.01, α = −0.4, β = 0.08, and
0
transmission bands to bew formed. Thus, power can be ǫ0 =0.3.
transmitted not only for frequencies in the linear band,
butalsointheotherwiseforbiddenfrequencyregions. As When losses are small, the appearance of more non-
it can be observed in Fig. 7, a number of transmission linear bands is possible, like the one seen just above the
bands have appeared due to the nonlinearity. The ar- linear band in Fig. 7, at f 1 GHz. That transmission
∼
ray thus becomes transparent in power transmission at bandis due to aninterplaybetween nonlinearityandge-
frequency intervals around the nonlinear resonances of a ometrical (lattice) resonances, and it is very sensitive to
single varactor-loadedSRR oscillator. That self-induced changesintheparametersofthemodel. Notethatinthe
transparency is a robust effect and survives with consid- density plots presented above the frequency is given in
erably increased losses, as it is shown in Fig. 8, where natural units, using the relation f =f Ω=0.9Ω GHz.
0
8
It is very illuminating to see the time evolution of the 0
current flowing in each SRR of the array for some fre-
quencies in the transmissionbands. We have chosentwo
different frequencies, one in the linear band and one the
)
m -50
high frequency band which is around the second har-
B
monic of the linear resonance frequency. The time evo-
d
lution of the currents in the first, fourth, seventh, and P(
tenth (last) SRR are shown for f = 0.888 GHz and
-100
f = 1.78 GHz in Figs. 9a and 9b, respectively. The
currentsoscillateperiodicallyinallSRRs, althoughwith
different amplitudes and relative phases with respect to
0 2 4 6 8 10
the driver. This figure suggests that power transmis- n
sion for frequencies in the forbidden band gap occurs
through the linear modes of the system. For frequen- FIG. 10: Power tranfer efficiency in different frequency re-
cies outside the linear and nonlinear bands, the currents gions, i.e.,averagepower(indBm)asafunctionofsitenum-
exhibit similar oscillations, althoughtheir amplitude be- ber n for an array ε0 = 0.3, λ = 0.01, γ = 0.001, N = 10,
comes vanishingly small with increasing site number n. and f = 0.888 GHz (black diamonds); f = 1.197 GHz (red
squares), f =1.78 GHz (green triangles).
The relative efficiency of power transmission in the two
the end of the extended array, and no reflection occurs.
2
The total number of SRRs is now N′ = N +N where
1
N =5,andthedissipationcoefficientγisnowafunction
1
n 0
i of the site number n, γ(n), given by
-2 (a) γ , 1 n N
γ(n)= γeµn , N≤<n≤ N′ , (34)
(cid:26) ≤
1
where µ = log(1/γ), with γ = 0.001 as before. For the
N1
arraywithn dependentdissipationcoefficient,atypical
−
averagepowerdensityplotisshowninFig. 11,wherethe
n 0
i power in each site is the average over a time interval of
5000 T . We observe, at least for the first N sites of the
0
(b) array,a pattern similar with those seen in Figs. 7 and 8.
-1
That is, the appearance of the linear and the nonlinear
0 5 1τ0 15 20 transmission bands at about the same frequency inter-
vals. We also observe that almost all of the transmitted
power is absorbed from the five last sites of the array.
FIG.9: Timedependenceofthecurrentin inthen−thSRR
Importantly, the narrow band coming from geometrical
for an array with N =10, λ=0.01, ǫ0 =0.3, γ =0.001, and
resonancesinFigs. 7and8,whichisaroundf =1GHz,
frequency (a) f = 0.888 GHz; (b) f = 1.78 GHz. The four
different curves are for the current in the n=1 SRR (black- appearsin Fig. 11as well,and moreoverit seems that it
solid), n = 4 SRR (red-dashed), n = 7 SRR (green-dotted), is capable of transmitting power to the end of the array.
n=10 SRR(blue-dot-dashed).
frequenciesusedinthe previousfigure,areshowninFig.
10, along with the power transmission for a specific fre- V. DISCUSSION AND CONCLUDING
quency(f =1.197GHz)outsidealllinearandnonlinear REMARKS
bands. It is remarkable that the transmission efficiency
is almostthe sameforfrequencies inthe twopass-bands, We have used a simple electric circuit model of in-
whilefortheotheronethetransmittedpowerpractically ductively coupled resistive-inductive-capacitive circuits
vanish at the fourth site. with voltage-dependent capacitance to investigate the
In the density plots above,the rightboundary wasac- transmission of power in a one-dimensional, end-driven,
tually a reflectingone,allowingthe formationofstation- nonlinear magnetic metamaterial comprised of varactor-
ary states in the array. We have also used a totally ab- loaded SRRs. We have used both a reflecting and an
sorbing boundary by adding five more sites at the right absorbingboundaryfortheendthatisnotdrivenbythe
end ofthe array,where the damping coefficientincreases external field. The importance of such discrete models
exponentially with increasing site number. Thus, the fordescribingthedynamicbehaviourofSRR-basedmag-
waves coming from the left are completely damped at netic metamaterials has been recently appreciated [27],
9
Their width apparently depends on the frequency, i.e.,
the bands at higher frequencies are significanlty wider
15 0 than those at lower frequencies.
Moreover, we have also observed the formation, ei-
-50
12 ther complete or partial, of another nonlinear pass-band
-100
at frequencies slightly above the linear band (at f
∼
9 -150 1 GHz). That band, which is rather narrow and can
n
-200 be observed both in Figs. 7 and 8 as well as in Fig. 11,
6 seem to depend sensitively on the model parameters λ,
-250
f, ε (for α and β fixed), and mainly on the loss coef-
0
-300
3 ficient γ. Comparing Figs. 7 and 8, which have been
-350 obtained for different loss coefficient γ =0.001 and 0.01,
0 0.5 1 1.5 2 -400 respectively, we see that the power transmission in that
f (GHz) band considerably decreases when the losses increase by
an order of magnitude. That type of nonlinear band
appears due to geometrical resonances of the standing
FIG.11: AveragepowerdensityplotPn(indBm)onthen−f waves that are formed in the array due to driving at
plane for a varactor-loaded SRR array with absorbing right one end. Geometrical resonances seem to offer an im-
boundary (see text), for λ = +0.01, ǫ0 = 0.297, γ = 0.001, portant mechanism for the generation of nonlinear pass-
and N′ =15 (N =10). bands,whichcanbeemployedforconstructingtransmis-
sion lines with banded power transmission spectra with
bands at desired frequency ranges. Similar conduct-free
althoughsimilardiscretemodelshavebeenemployedear- transmission lines could be formed by superconducting
lierforthestudyoflocalizedexcitationsinthosesystems magnetic metamaterials, where the basic structural ele-
[16, 35] ment(theSRR)isreplacedbyitsdirectsuperconducting
In the present work, we have focused on the trans- analogue, the rf SQUID [17, 19]. The rf SQUID, where
mission of power along the array, and especially that in the acronym stands for radio-frequency superconductive
the nonlinear regime, which results in the formation of quantum interference device, is just a superconducting
nonlinear pass-bands for frequencies close to the reso- ring interrupted by a Josephson junction, which makes
nances of a single varactor-loadedSRR that is described it an intrinsically nonlinear element. A one-dimensional
by the approximate model equation (13). The approx- arrayof rf SQUIDs coupled through their mutual induc-
imate model holds fairly well for relatively low driving tances forms a nonlinear magnetoinductive transmission
amplitudes,forwhichthedissipativediodecurrentofthe line[19]whosepowertransmissionpropertiesisamatter
varactor can be neglected. It seems, thus that the prop- of future work.
erties of the individual elements of the array predomi-
nantly determine its transmission properties, at least in
the case of weakmagnetoinductive coupling investigated
Acknowledgments
in the present work. The numerical results demonstrate
that the nonlinear pass-bands transmit power with ef-
ficiently comparable to that of the linear band. For the This workwassupportedin partby the EuropeanOffice
parametersetsusedinthiswork,whichareclosetothose ofAerospaceResearchandDevelopment, AFSORaward
in [50] for a single varactor-loaded SRR, there are three FA8655-10-1-3039, and by the EURYI and MEXT-CT-
nonlinear bands, both above and below the linear band. 2006-039047.
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