IEEE MTT-V056-I02 (2008-02)

308 Pages·2008·40.746 MB·English

by  FANTOMASPING

FEBRUARY2008 VOLUME56 NUMBER2 IETMAB (ISSN0018-9480)
PAPERS
SmartAntennas,PhasedArrays,andRadars
Millimeter-Wave-RadarSensorBasedonaTransceiverArrayforAutomotiveApplications ................................
.......................................................................... M.Steinhauer,H.-O.Ruoß,H.Irion,andW.Menzel 261
AMonolithicPhasedArrayUsing3-bitDistributedRFMEMSPhaseShifters...............................................
...........................................................................K.Topalli,O.A.Civi,S.Demir,S.Koc,andT.Akin 270
24-GHzFrequency-ModulationContinuous-WaveRadarFront-EndSystem-on-Substrate ................ Z.LiandK.Wu 278
ActiveCircuits,SemiconductorDevices,andICs
ASubthresholdLow-NoiseAmplifierOptimizedforUltra-Low-PowerApplicationsintheISMBand ....................
..................................................................... A.V.Do,C.C.Boon,M.A.Do,K.S.Yeo,andA.Cabuk 286
-and -BandCompactOctaveBandwidth4-bitMMICPhaseShifters ........................ I.J. BahlandD.Conway 293
A2-GHzGaAsHBTRFPulsewidthModulator .....................................................M.NielsenandT.Larsen 300
ACompactHighlyReconfigurableCMOSMMICDirectionalCoupler .......................................................
.......................................................................... M.A.Y.Abdalla,K.Phang,andG.V.Eleftheriades 305
FastEstimationofSpectralSpreadinginGSMOPLLTransmittersBasedonFoldingEffectsAnalysisinQuadraturePhase
Modulator ................................................................................................................H. Shin 320
AnalysisofaFullyMatchedSaturatedDohertyAmplifierWithExcellentEfficiency ........................................
............................................................. J.Kim,J.Moon,Y.Y.Woo,S.Hong,I.Kim,J.Kim,andB.Kim 328
16.6-and28-GHzFullyIntegratedCMOSRFSwitchesWithImprovedBodyFloating .....................................
................................................................................... Q.Li,Y.P.Zhang,K.S.Yeo,andW.M.Lim 339
SignalGeneration,FrequencyConversion,andControl
ATriple-TunedUltra-WidebandVCO ............................................................................................
....................... M.Tsuru,K.Kawakami,K.Tajima,K.Miyamoto,M.Nakane,K.Itoh,M.Miyazaki,andY.Isota 346
AnalysisandDesignofaDouble-QuadratureCMOSVCOforSubharmonicMixingat -Band .........................
.............................................................................A.Mazzanti,E.Sacchi,P.Andreani,andF.Svelto 355
Millimeter-WaveandTerahertzTechnologies
SchottkyBarrierDiodeCircuitsinSiliconforFutureMillimeter-WaveandTerahertzApplications ........................
...................................................... U.R.Pfeiffer,C.Mishra,R.M.Rassel,S.Pinkett,andS.K.Reynolds 364
(ContentsContinuedonBackCover)
(ContentsContinuedfromFrontCover)
WirelessCommunicationSystems
Multi-LookupTableFPGAImplementationofanAdaptiveDigitalPredistorterforLinearizingRFPowerAmplifiersWith
MemoryEffects .....................................P.L.Gilabert,A.Cesari,G.Montoro,E.Bertran,andJ.-M.Dilhac 372
ANewWidebandAdaptiveDigitalPredistortionTechniqueEmployingFeedbackLinearization...........................
......................................................................................... J.Kim,Y.Y.Woo,J.Moon,andB.Kim 385
CADAlgorithmsandNumericalTechniques
Phase-NoiseAnalysisofInjection-LockedOscillatorsandAnalogFrequencyDividers .....................................
.............................................................................. F.Ramírez,M.Pontón,S.Sancho,andA.Suárez 393
EfficientAlgorithmsforCrank–Nicolson-BasedFinite-DifferenceTime-DomainMethods ...................... E.L.Tan 408
ExactEquivalentStraightWaveguideModelforBentandTwistedWaveguides ............................. D.M. Shyroki 414
FiltersandMuliplexers
ModelingandOptimizationofCompactMicrowaveBandpassFilters .............. M. Bekheit,S.Amari,andW.Menzel 420
ACompactOpen-LoopFilterWithMixedElectricandMagneticCoupling ...................... Q.-X.ChuandH.Wang 431
MiniaturizedHexagonalStepped-ImpedanceResonatorsandTheirApplicationstoFilters .................................
................................................................................ R.-J.Mao,X.-H.Tang,L.Wang,andG.-H.Du 440
RFAmplitudeandPhase-NoiseReductionofanOpticalLinkandanOpto-ElectronicOscillator ..........................
............................................................................................D.Eliyahu,D.Seidel,andL.Maleki 449
ASeriesSolutionfortheSingle-ModeSynthesisProblemBasedontheCoupled-ModeTheory ...........................
............................................................ I.Arnedo,M.A.G.Laso,F.Falcone,D.Benito,andT.Lopetegi 457
Packaging,Interconnects,MCMs,Hybrids,andPassiveCircuitElements
DesignConsiderationsofMiniaturizedLeastDispersivePeriodicSlow-WaveStructures .... C.ZhouandH.Y.D.Yang 467
VerticalTopologiesofMiniatureMultispiralStackedInductors ................................................................
................................................................W.-Y.Yin,J.-Y.Xie,K.Kang,J.Shi,J.-F.Mao,andX.-W.Sun 475
ANovelApproachtotheDesignandImplementationofDual-BandPowerDivider ........ K.-K.M.ChengandC.Law 487
InstrumentationandMeasurementTechniques
Six-PortReflectometerBasedonModifiedHybridCouplers ...........................................J.J.YaoandS.P.Yeo 493
Microwave(8–50GHz)CharacterizationofMultiwalledCarbonNanotubePapersUsingRectangularWaveguides......
....................................................................................................L.Wang,R.Zhou,andH.Xin 499
SynthesisofaWidebandMultiprobeReflectometer .............................................................................
................................................. B.M.Kats,A.A.Lvov,V.P.Meschanov,E.M.Shatalov,andL.V.Shikova 507
Comb-GeneratorCharacterization .............................. H.C.Reader,D.F.Williams,P.D.Hale,andT.S.Clement 515
MEMSandAcousticWaveComponents
Wafer-ScalePackagedRFMicroelectromechanicalSwitches...................................................................
............................................................. J.Muldavin,C.O.Bozler,S.Rabe,P.W.Wyatt,andC.L.Keast 522
NovelHigh- MEMSCurled-PlateVariableCapacitorsFabricatedin0.35- mCMOSTechnology.......................
............................................................................. M.Bakri-Kassem,S.Fouladi,andR.R.Mansour 530
Photonic Generation of Chirped Millimeter-Wave Pulses Based on Nonlinear Frequency-to-Time Mapping in a
NonlinearlyChirpedFiberBraggGrating............................................................... C.WangandJ.Yao 542
Biological,Imaging,andMedicalApplications
FDTDCalculationsofSpecificAbsorptionRateinFetusCausedbyElectromagneticWavesFromMobileRadioTerminal
UsingPregnantWomanModel..................................................................................................
...................................... T.Togashi,T.Nagaoka,S.Kikuchi,K.Saito,S.Watanabe,M.Takahashi,andK.Ito 554
InformationforAuthors ............................................................................................................ 560
CALLSFORPAPERS
SpecialIssueonRFIDHardwareandIntegrationTechnologies ................................................................ 561
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AMIRMORTAZAWI AssociateEditors
Univ. of Michigan
AnnArbor,MI48109-2122USA DANIELDEZUTTER JENSHAN LIN ZOYA POPOVIC KE-LI WU
Phone:[phone] Universiteit Gent Univ. of Florida Univ.ofColorado,Boulder ChineseUniv.ofHongKong
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email:[email protected] email:[email protected] email:[email protected] email:[email protected] email:[email protected]
DYLAN WILLIAMS YOUNGWOO KWON JOSÉ PEDRO SANJAY RAMAN RUEY-BEEI WU
NIST Seoul Nat. Univ. Univ. of Aveiro VirginiaPolytech.Inst.andStateUniv. NationalTaiwanUniv.
Boulder,CO80305USA Korea Portugal USA Taiwan, R.O.C.
Phone:[phone] email:[email protected] email:jcp.mtted.av.it.pt email:[email protected] email:[email protected]
Fax:[phone] RICHARD SNYDER ALEXANDERYAKOVLEV
email:[email protected] RSMicrowaveCompany Univ. of Mississippi
USA USA
email:[email protected] email:[email protected]
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DigitalObjectIdentifier10.1109/TMTT.2008.917703
IEEETRANSACTIONSONMICROWAVETHEORYANDTECHNIQUES,VOL.56,NO.2,FEBRUARY2008 261
Millimeter-Wave-Radar Sensor Based on a
Transceiver Array for Automotive Applications
MatthiasSteinhauer, Member, IEEE, Hans-Oliver Ruoß, Member, IEEE, Hans Irion, and
Wolfgang Menzel, Fellow, IEEE
Abstract—Thispaperdescribesanewradarsensorarchitecture
comprising an array of transceiver modules. Target applications
of the sensor are automotive driver assistance systems. In con-
junction with a monolithic integration of each transceiver, the
conceptoffersthepossibilityforacost-effectiverealizationofdig-
ital-beamforming radar sensors at millimeter-wave frequencies.
A modulation sequence is investigated based on simultaneously
transmittedfrequency-modulatedcontinuous-wavesignals,which
are separated by frequency multiplexing. Appropriate signal
processingfortheestimationofrange,speed,andazimuthangle
inmultipleobjectsituationsispresented.
Experimentalresultswithaneight-channelradarsensorinthe
76–77-GHzfrequencybandarepresented,whichdemonstratethe
feasibilityoftheproposedarchitectureandshowtheperformance
ofthemodulationsequenceandsignalprocessing.
Index Terms—Array, automotive radar, digital beamforming
(DBF), frequency-modulated continuous wave (FMCW), trans-
ceiver.
Fig.1. BlockdiagramofatypicalDBFradarsensorfrontend.
I. INTRODUCTION
MILLIMETER-WAVE (MMW) radar is an important main.Atypicaldigitalbeamforming(DBF)frontendarchitec-
ture is shown in Fig. 1. The main circuit blocks are a signal
sensing principle for automotive driver assistance sys-
source, which feeds a single transmitantenna, and an array of
tems. The main automotive application of radar sensors today
receivers where a part of the transmit signal is used as a local
is adaptive cruise control, which is based on forward-looking
oscillatorsignaltodown-converttheRFsignalstobaseband.
long-rangesensorsinthe76–77-GHzfrequencybandcovering
The cost associated with this architecture has prohibited its
a range up to 150 m and an azimuth field of view of 8 .
implementationforautomotiveradarsensorsthusfar.Different
Accordingtotheevolutionofdriverassistancesystemstowards
concepts based on switching in the transmit or receive paths
active safety systems, the radar sensors will also be employed
havebeen proposed toreduce the required numberof receiver
for functions such as collision warning, brake assistance, or
channels [2]–[7]. However, switching operation at MMW fre-
automatic emergency braking. These applications demand
quencies of 76 GHz introduces significant losses, which are
a higher detection performance compared to current radar
disadvantageous in the receive path due to an increased noise
sensors. The azimuthal angular resolution has to especially be
figureandinthetransmitpathduetothelimitedtransmitpower
improved significantly to values below 2 to achieve a better
ofmonolithicsignalsources.Switchingtypicallyalsorequires
objectseparation[1],[2].Simultaneously,alargerfieldofview
anincreasedbandwidthinthebaseband,whichadditionallyin-
is required.
creasesnoisepowerand,therefore,reducessystemsensitivity.
Asuitablesensorarchitectureforimprovedangulardetection
A different approach to reduce the cost of an array sensor
performanceisanantennaarraywithbeamformingorhigh-res-
architecture is pursued with the sensor concept proposed in
olutiondirectionofarrival(DOA)estimationinthedigitaldo-
this paper. The proposed architecture comprises an array of
transmit–receivemodules,whichcanbecost-effectivelyimple-
mented by monolithically integrating each transceiver module
ManuscriptreceivedFebruary1,2007;revisedAugust28,2007.
M. Steinhauer is with Chassis Systems Control, Robert Bosch GmbH, includingantennastructures,therebyavoidinginterconnectsat
D-74232Abstatt,Germany([email protected]). MMWfrequencies.
H.-O.RuoßiswithAutomotiveElectronics,RobertBoschGmbH,D-72703
Reutlingen,Germany([email protected]).
H. Irion is with Gasoline Systems, Robert Bosch GmbH, D-71701 II. SENSORCONCEPT
Schwieberdingen,Germany([email protected]). A block diagram of the radar sensor frontend is depicted
W.MenzeliswiththeInstituteofMicrowaveTechniques,UniversityofUlm,
in Fig. 2. The channels of the array are realized as identical
D-89069Ulm,Germany([email protected]).
DigitalObjectIdentifier10.1109/TMTT.2007.914635 transmit–receivemodulesinsteadofreceiversinaconventional
0018-9480/$25.00©2008IEEE
262 IEEETRANSACTIONSONMICROWAVETHEORYANDTECHNIQUES,VOL.56,NO.2,FEBRUARY2008
Fig.2. Principleblockdiagramofproposedtransceiverarrayfrontend.
Fig.3. Timesectionofthetransmitsignalswithfrequencymultiplexbetween
array frontend architecture. This arrangement is advantageous thedifferenttransceivermodules.
from a cost perspective in combination with a monolithic
integration of each transceiver including antenna structures.
The monolithic integration of the transceiver modules allows be achieved, and no additional signal processing effort is re-
anassemblywithouttheneedofinterconnectsandwirebonds quiredtoeliminatepotentialambiguitiesresultingfromaliasing
in the carrier frequency range of 76 GHz as no MMW signal effects.
has to be distributed between the transceivers. Therefore, the TheFMCWtransmitsignalofthe thsourcecanbedescribed
assembly is simplified, and a standard printed circuit board as
substratematerialcanbeused.Themultipledeploymentofone
monolithicmicrowaveintegratedcircuit(MMIC)alsoleadsto
(1)
ahighmodularityofthesensorconcept.
In principle, all basic radar modulation signals as FMCW,
where and arethebandwidthanddurationofthemodu-
pulse Doppler, or pseudonoise phase coding can be used as
lation,respectively, isthereferencecarrierfrequency, is
transmitsignalsofthetransceivermodules.TheFMCWmodu-
thefrequencyshiftbetweenthecarrierfrequenciesofadjacent
lationischosenherebecauseofthelowcomplexityrequiredfor
modules,and istheconstantcarriersignalphaseofthe th
thetransceivermodulesandtheefficientuseofavailablesignal
source.Areceivedechosignalofthe thmodule,resultingfrom
power.Theavailabilityofindependentsignalsourcesforeach
areflectionofitsowntransmitsignalatamovingobject,isan
arraymodulefacilitatestheformationofdifferenttransmit–re-
attenuatedandtime-delayedreplicaofthetransmitsignalwith
ceive configurations of the array. A favorable configuration is
aDopplershiftofthecarrierfrequency
themonostatictransmit–receivemodeofeachtransceiverwith
homodyne conversion of the received signals, as depicted in
Fig. 2. The main advantage of this configuration is the elim-
ination of the carrier signal phases by the homodyne mixing (2)
process whereby the need for exact phase synchronization of
thesourcesisavoided.Aprerequisiteofthismodeisthedecou- where istheattenuation, istheobjectrangetothereference
plingofthemodulestoavoidtransmitterinterference. module, istheobjectspeed,and isthefree-spacepropaga-
The modulation schemes developed for switched FMCW tionvelocity.
arrays represent a time-domain multiplex of the channels and The baseband signal results from the multiplication of
could,therefore,be appliedtothetransceiverarray.Examples transmit and receive signal. For further analysis, only the
are the processing of sequential fast frequency-modulated phase of the resulting signal is relevant. Taking the module
continuous-wave(FMCW)sweeps[6],[7]orthefragmentation at the left edge of the array as the reference, the phase of the
ofasingleFMCWsweepuponthedifferentchannels[2]. basebandsignalofthe thmodulecanbeexpressedaftersome
Here,adifferentmodulationschemeisproposed,makinguse approximationsas
of the available signal sources in each channel. The time–fre-
quencyrelationofthetransmitsignalsofthedifferent sources
is depicted in Fig. 3. All sources are transmitting simultane-
ously,andtransmitterinterferenceisavoidedbyfrequencymul-
(3)
tiplexing of the carrier frequencies. Advantages of this mod-
ulation scheme are the avoidance of switching operations and
theminimizedcycletimeduetothesimultaneoustransmission. where is theazimuthangleoftheincomingsignal, isthe
Bythepreventionofswitching,abettersystemsensitivitycan free-spacewavelength,and istheelementspacingofthearray.
STEINHAUERetal.:MMW-RADARSENSORBASEDONTRANSCEIVERARRAYFORAUTOMOTIVEAPPLICATIONS 263
Basebandsignalsresultingfromtransmitsignalsofallother
modules can be suppressed by a low-pass filter as they are
shiftedbythetransmitfrequencydifference .
Thetime-dependentphasetermin(3)representsthebaseband
frequency, which is dependent on distance and velocity of an
object
(4)
The progression of the time-independent baseband signal
phasealongthearraycorrespondstothephasesofthesteering
vector components and is dependent on the angle of arrival Fig.4. Modulationsequence.
of the incoming signal. Additionally the phase progression
is dependent on the distance of the reflecting object. This
basebandfrequencyaccordingto(4).Acommonapproachtore-
relation is caused by the carrier frequency shift between the
solvetheambiguityofthebasebandfrequencyistheprocessing
transceiver modules. By evaluating time-shifted sections of
ofseveralFMCWsweepswithdifferentmodulationparameters,
the baseband signals, an effective time and frequency shift is
e.g.,atriangularwaveformwithanascendingandadescending
realized between the FMCW signals of adjacent modules, as
frequency sweep [8]. This yields two linear equations, which
depicted in Fig. 3. For the effective time and frequency shift
canbesolvedunambiguouslyonlyforasingleobject.Inmul-
andthephysicalfrequencyshift,thefollowingrelationholds:
tipleobjectsituationswith objects,thisleadsto pos-
sibleobjects,andtheso-calledghosttargetshavetoberesolved
(5)
byfurtherFMCWsweeps.Thisresultsinahighprocessingef-
where and are the effective frequency and time shifts, fort,andtheriskofunresolvedghosttargetsremains.
respectively. This leads to a total phase progression along the The unambiguous estimation of distance and speed is pos-
array, which is dependent on azimuth angle, distance, and ve- sible by the utilization of frequency- or time-shifted FMCW
locityofanobject sweepsbyusingtheadditionalphaseinformation[9],[10].The
underlying principle is the dependency of the baseband phase
(6) difference from distance and velocity when applying a time
or frequency shift. With the system concept and modulation
schemeproposedhere,thevirtualshiftoftheestimatedazimuth
The result is a virtual shift of the detected azimuth angle ac-
anglefromtheactualvaluecanbedeployedtoextracttherele-
cording to
vantphaseinformation.
Asuitablemodulationsequenceisrepresentedbythefirsttwo
(7)
FMCWsweepsdepictedinFig.4.Inthefirststepofthesignal
processing,basebandfrequenciesandassociatedazimuthangles
withtheestimatedazimuthangle ,therealazimuthangle ,
areestimatedforanyreflectedsignalwithineachsweep.Dueto
andtheobjectvelocityanddistance and .
the identical sweep rate of each channel, equal baseband fre-
Furthermore, the phase lags between adjacent modules are
quenciesresultforeachchannel.Thebasebandfrequenciescan
doubledcomparedtoaconventionalreceiverarraywithasingle
be efficiently estimated by a fast Fourier transform (FFT) and
transmit source. This results from the simultaneous variation
subsequent peak detection. The associated azimuth angles are
oftransmitterand receiverlocationduetothe monostaticpro-
estimated by the application of DBF algorithms to the ampli-
cessing of the respective transmit and receive signals as the
tudeandphaseinformationinthefrequencydomain.
phaselagsoftransmitandreceivepathareadded[3]–[5].The
For the ease of calculation, the wavenumber in direction of
additionofthephaselagsis,inthecurrentcase,equivalenttoa
the array
doublingofthephysicalapertureofthearray.Therefore,are-
quiredangularresolutioncanbeachievedwithasmallerphys-
(9)
icalaperture.Asthenumberofelementsiskeptconstant,theun-
ambiguousangularrangedeterminedbythedistanceofgrating
isusedinsteadoftheazimuthangle,assumingthe -axisisori-
lobesofthearrayfactorisreducedto
entedalongthearray.Rewriting(7)resultsin
(8)
(10)
III. MODULATIONSEQUENCEANDSIGNALPROCESSING Togetherwiththebasebandfrequency,oneFMCWsweepre-
The additional dependence of the estimated azimuth angle sultsintwolinearindependentequations,whichareconnected
fromobjectdistanceandspeedaccordingto(7)canbedeployed for each object, containing and . The equations cannot be
fortheunambiguousestimationofdistanceandspeed.Adraw- directly solved, as after (10), the estimated wavenumber
backoftheFMCWmodulationisthecouplingof and inthe isalsodependentontheunknownactualwavenumber .Two
264 IEEETRANSACTIONSONMICROWAVETHEORYANDTECHNIQUES,VOL.56,NO.2,FEBRUARY2008
consecutive FMCW sweeps according to the first two sweeps TABLEI
in Fig. 4 can be used to eliminate , as its value can be as- ARRAY, MODULATION, AND SIGNAL PARAMETERS
DEPLOYEDFORVARIANCECALCULATION
sumedtostayconstantduringthemeasurementcycle.Usinga
different carrier frequency shift between adjacent transceivers
in each sweep, two linearindependent equations result, which
canbesubtractedtoyield
(11)
From(11)and(4), and canbedirectlyestimatedas
(12)
(13)
The actual azimuth angle can subsequently be calculated
from(7)usingtheestimateddistanceandvelocity.
In general,the achievableaccuracy ofdistance and velocity
estimationusing(12)and(13)islowercomparedtotheuseof
twoindependentbasebandfrequencies.Thisisduetothelower
The estimation accuracy is dependent on the variance of fre-
accuracy of angle or phase estimation compared to frequency
quency estimation and angle estimation , respectively.
estimation. Therefore, if a higher accuracy is required, a third
Forthe modulationparameters andunambiguous distanceand
FMCW sweep can be used, as depicted in Fig. 4. Using the
velocityrangeslistedinTableI,the minimumstandarddevia-
resultingbasebandfrequency,theestimationaccuracyfromthe
tionisachievedformaximumadmissibleeffectivetimeshiftand
firststepaccordingto(12)and(13)isimproved.
zeroeffectivefrequencyshiftbetweenadjacenttransceiversig-
A. UnambiguousRange nals.Here,themaximumadmissibleeffectivetimeshiftequals
safter(15).
The phase difference or angle difference, respectively, in-
Using an FFT of length 512 with subsequent maximum de-
volvedinthedistanceandvelocityestimationaccordingto(12)
tectionandcenterofgravityestimation,astandarddeviationfor
and (13) is ambiguous. The unambiguous phase range corre-
the frequency estimation of approximately can
sponds to , where is the phase difference
beachievedforasignaloflength .Asapracticalestimation
betweenthebasebandsignalsofadjacenttransceivers.Therel-
accuracy for estimation of the azimuth angle, a standard devi-
evantphasedifferencebetweenthebasebandsignalsoftwoad-
ation of corresponding to a standard deviation of
jacenttransceiverscanbederivedfrom(11)as
m is assumed here, which is achievedwith the
experimentalradarsensordescribedinSectionIV.Theresulting
(14) standarddeviationsfortheestimationof and can,therefore,
becalculatedas
Considering the unambiguous phase range, the effective time
andfrequencyshiftsforunambiguousdistanceandvelocityes- m (18)
timationmustsatisfythefollowingrelation: m/s (19)
(15) It has to be considered that deviations of the nominal fre-
quencyoffsetbetween the carrierfrequencies lead toa bias in
theestimationofdistanceandvelocity.Asynchronizationofthe
wheretheunambiguousdistancerangeisassumedas
carrierfrequenciesis,therefore,requiredtoachievetheestima-
andtheunambiguousvelocityrangeas .
tionaccuracyderivedabove.
B. EstimationAccuracy
Thevarianceoftheestimationof and using(12)and(13) IV. EXPERIMENTALRADARSENSOR
canbederivedbyerrorpropagationas
A prototypical radar sensor for the 76–77-GHz frequency
range has been realized to investigate the feasibility of the
(16) system concept with emphasis on the modulation sequence
and signal-processing scheme. A block diagram of the sensor
is depicted in Fig. 5. The sensor comprises eight parallel
transmit–receive circuits, which are realized in a hybrid as-
(17) sembly of MMICs and discrete components on an organic
STEINHAUERetal.:MMW-RADARSENSORBASEDONTRANSCEIVERARRAYFORAUTOMOTIVEAPPLICATIONS 265
Fig.6. RFcircuitboard.
Fig.5. Blockdiagramofexperimentalradarsensorconsistingofeightseparate
transmit–receivemodules.
A photograph of the RF frontend is depicted in Fig. 6. In
high-frequency substrate. A hybrid assembly on a high-fre- thecenter,theMMICsforthesignalsourcesarerecognizable.
quency laminate is used, as no fully monolithic transceiver Theoutputsignalsaretransferredtotheantennacircuitsbymi-
modules were available yet. The essential properties of the crostriplineswhoselengthvariesbetweenthetransceiversdue
sensor concept can be evaluated independent from the as- toassemblyconstraints.Thepolyrodantennasarevisibleatthe
sembly technology. In parallel, a development of monolithic topofthepicture.
transceivermodulesforMMWfrequenciesisongoing.Inafirst The attenuation of the transmission lines between signal
step,thefeasibilityofkeycomponentsasoscillatorsandmixers source and transfer mixers varies from 1.5 to 5 dB due to the
forfrequenciesbeyond100GHzisshowninanSiGe–BiCMOS differentlinelengths.Theinsertionlossofthetransmitpathof
technology[11]. the mixer amounts to 3 dB. With the typical output power of
Atthecurrentprototypicsensor,MMICoscillatorsatanom- thefrequencydoublerof18dBmfrom76to77GHz,thesignal
inalfrequencyof38GHzareusedassignalsources.Foreach powerattheantennafeedpointscanbeestimatedtovaryfrom
channel, a subsequent frequency doubler and power amplifier 10to13.5dBmamongthechannels.
MMIC transfers the oscillator signal to the target frequency Thelow-frequencycircuitsasPLLapplication-specificinte-
range and provides a typical output power of 18 dBm. The gratedcircuits(ASICs),basebandamplifiersandfiltersareas-
mixers are realized as transfer mixers consisting of a single sembledonanFR4substrate,whichislaminatedtothebackside
series diode. A part of the source signal is coupled with a oftheRFsubstrate.Forthesensorcontrolanddataacquisition,
directional coupler and a bandpass filter to a harmonic mixer, aseparateboardwithaMotorolaMPC5200microprocessoris
where it is converted to an intermediate frequency of 2 GHz. used. The microprocessor is controlled from a PC using Lab-
A 18.6-GHz dielectric resonator oscillator provides the local View,andacontrollerareanetwork(CAN)interfaceisusedfor
oscillator signal for all eight channels. Its signal is distributed communication.
to the eight subharmonic mixers by a three-stage Wilkinson The bandwidth of the system is limited by the maximum
power-divider network. The nonlinear component of the sub- sweepbandwidthofthePLL-ASICtoapproximately500MHz.
harmonicmixersisasinglediodeofthesametype,asitisused
forthereceivingmixers.A15-MHzquartzoscillatorisusedas
a common reference source for the phase locked-loop (PLL) V. MEASUREMENTRESULTS
circuits.
The antennas consist of half-wavelength microstrip patches
A. PhaseNoiseofTransmitSignals
asradiatingelementswithdielectricrodsontop.Thedielectric
rods have a lower cylindric and an upper conical section with For characterization of the phase noise of the sources, the
an overall length of approximately 8 mm. The geometric pa- transmitted signals at a fixed frequency of approximately
rameters as length, diameter, and taper of the conical section 76.5GHzarereceivedbymeansofahornantennaclosetothe
have been optimized with the help of a full-wave electromag- sensor.A40-GHzspectrumanalyzerwithanexternal -band
neticsimulationtoachieveanefficientilluminationofthesub- downconverterisusedtomeasurethepowerspectrum.
sequentlens.Acylindricdielectriclensisdeployedtoobtaina ThepowerspectrumofonechannelisdepictedinFig.7.The
narrowelevationbeamandanincreasedantennagain.Theinflu- phasenoisecanbeestimatedtoapproximately 80dBc/Hzat
enceontheazimuthradiationcharacteristicisintentionallylow 100-kHz offset from the carrier. At this measurement, the fre-
toachievethelargestpossiblebeamoverlapoftheeightchan- quencyspacingbetweenthetransmitsignalsofdifferentchan-
nels. A spacing of the patch antennas of one free-spacewave- nelsis18MHz.Withdecreasingfrequencyspacing,thephase
lengthischosen. noiseincreasesduetoparasiticcouplingbetweenthePLLs.For
266 IEEETRANSACTIONSONMICROWAVETHEORYANDTECHNIQUES,VOL.56,NO.2,FEBRUARY2008
Fig.7. Powerspectrumoftransmitsignalofonetransceiverchannel. Fig. 9. Normalized array pattern for reflector at 0 with 25-dB Chebyshev
weightingappliedtothebasebandsignals.
The azimuth pattern has a sector-like form with a 6-dB
two-way beamwidth, corresponding to the one-way 3-dB
beamwidth of approximately 52 , and a steep transition from
themainlobetothesideloberegion.Theelevationpatternhas
a6-dBtwo-waybeamwidthof4 duetothedielectriclens.The
highestsidelobesareapproximately30dBbelowthemainlobe
forthetwo-waypattern.
2) ArrayPattern: Thearraypatternresultingafterarraycal-
ibrationisdepictedinFig.9.AChebyshevwindowwith25-dB
sidelobesuppressionisappliedtothesignalsoftheeightchan-
nels.Arraycalibrationhasbeenperformedusingthealgorithm
describedin[12].The3-dBbeamwidthofthemainlobeis3.9
andthefirstzeroofthearraypatternislocatedat4.6 .Grating
lobesoccurat 30 and 90 ,whichresultfromtheeffective
spacingoftheantennasoftwowavelengthsduetothesynthetic
doublingofthephysicalspacingwhenmonostatictransmit–re-
ceiveprocessingisapplied.TheChebyshevweightingiseffec-
Fig.8. Normalizedtwo-waysingle-channelazimuthandelevationradiation tiveonlyforsidelobesbeyond 10 .Theclose-insidelobesare
pattern.
limitedtoapproximately20dBbelowthemainlobeduetocali-
brationimperfections.Ithastobenotedthatthedynamicrange
withrespecttointerferingsignalsbeyond 30 issubstantially
all measurement results, presented below, the same frequency
enhancedduetothesingle-channelpattern.
spacingof18MHzischosen.
C. RangeandVelocityEstimation
B. AntennaRadiationPattern
1) StaticObjects: Theestimationofrangeandvelocityusing
1) Single-ChannelPattern: Withmonostaticoperationofthe the principle modulation scheme, as presented in Section III,
transceivermodules,theeffectiveantennacharacteristicscorre- hasbeeninvestigatedexperimentallyforstaticanddynamicob-
spond to the two-way radiation pattern. This characteristic is jects.Thestaticmeasurementsetupconsistsofacornerreflector
measured here for all eighttransceiversby analyzing the peak in a distance of approximately 6 m to the sensor. The size of
power of the FMCW baseband signals resulting from the re- thereflectorisvariedtoanalyzetheinfluenceofsignal-to-noise
flected power of a corner reflector in a fixed distance of 6 m. ratio.Thewholesetupisplacedinananechoicchamber.Sweep
Themeasuredtwo-wayazimuthandelevationcharacteristicsof bandwidth and duration of the FMCW sweep are chosen as
onechannelaredepictedinFig.8.Theantennacharacteristicis MHzand msforalleightsignalsources.
normalizedtothemaximumbasebandsignalpower.Asimula- Thecarrierfrequencydifferencebetweenthetransmitsignalsof
tion of the antenna system including losses shows an absolute adjacentmodulesischosenlargeenoughtoallowadecoupling
antennagainof20dB. ofthetransceiversignalsbyfilteringinthebaseband.
STEINHAUERetal.:MMW-RADARSENSORBASEDONTRANSCEIVERARRAYFORAUTOMOTIVEAPPLICATIONS 267
Fig.10. StandarddeviationofthedirectestimationofRfrom(12)indepen- Fig.12. EstimateddistanceofanapproachingtargetforestimationofRfrom
denceofeffectivefrequencyshift(cid:14)f andsignal-to-noiseratio. (12)andusingtwoindependentbasebandfrequencies.
,whicharerealisticvaluesfortheexperimental
sensor.
The measured curves show the principal reciprocal rela-
tionship of the standard deviation to the effective time and
frequency shifts according to (16). The standard deviation
decreases, as expected, with increasing signal-to-noise ratio.
Consideringtherequiredunambiguousdistancerangeof200m
and unambiguous velocity range of 250 km/h, the maximum
permissible time shift is s and the maximum fre-
quency shift is kHz. For a signal-to-noise ratio of
20dB,theresultingstandarddeviationforatimeshiftof14 s
is cm. The corresponding standard deviation for the
velocityestimationyields m/s.
2) Dynamic Objects: The estimation of and using the
proposed signal-processing scheme utilizes differential phase
informationfromtwoconsecutiveFMCWsweeps,asdepicted
in the modulation cycle in Fig. 4. A prerequisite for unbiased
rangeandvelocityestimationisthattheazimuthangleofatarget
Fig.11. StandarddeviationofthedirectestimationofRfrom(12)indepen-
denceofeffectivetimeshift(cid:14)tandsignal-to-noiseratio. shows a negligible change between the two measurements as
theazimuthangleisassumedasequal.Toconfirmthevalidity
of this assumption, measurements with dynamic objects have
Thestandarddeviationforthedistanceestimationisdepicted been conducted. In Fig. 12, the results of distance estimation
inFigs.10and11fordifferenteffectivefrequencyshiftsanddif- overtimeforameasurementsetupwithastaticsensorandacar
ferenteffectivetimeshifts,respectively.Themeasuredstandard approachingthesensoraredepicted.Duringthemeasurement,
deviationforvelocityestimationisnotdepicted.Itshowsqual- thecarisacceleratingfrom10to30km/h.Theresultsofdirect
itativelyequalresultsduetothecouplingof and according estimationof and areshownforaneffectivefrequencyshift
to (4). The standard deviation is estimated from the results of of only kHz and for an effective time shift of only
15measurementsforeachparameter.Here,timeandfrequency s.Additionallytheresultsofthefollowingdistanceand
shift denote the difference of the effective time or frequency velocityestimationaredepictedusingthebasebandfrequencies
shiftofadjacenttransmitsignalsbetweenthefirsttwocommen- of the second FMCW sweep and a third sweep with different
surable frequency sweeps according to Fig. 4. Physically, the frequencyslope.Theresultsshowthatthedistanceandvelocity
frequencyshiftisvaried.Theeffectivetimeandfrequencyshifts estimation deploying differential phase information is reliably
areappliedbyanalyzingappropriatelytime-shiftedsectionsof applicabletodynamicscenarios.
thebasebandsignals.Additionally,thetheoreticalcurvesforthe
D. AzimuthAngleEstimation
standarddeviationafter(16)atasignal-to-noiseratioof20dB
are depicted, assuming a standard deviation of angle estima- Fortheevaluationofangularresolutionintheazimuthdirec-
tion of and of baseband frequency estimation of tion, a measurement setup with two equidistant corner reflec-